Tuesday, June 9, 2020

An Alternate Method for Characterizing Baluns and Common-Mode Chokes

Not long ago the NanoVNA group had a thread discussing the Nooelec 1:9  Balun and how to characterize it using the NanoVNA.

The topic was interesting, and so I thought I'd make my own measurements.  Specifically, I wanted to measure "Operating Power Gain" (also known as 'Gp'), which can be a truer measure of balun power loss, compared to Insertion Loss.  And I also wanted to check the balun's Common-Mode impedance.

A common technique for measuring balun loss is to connect two identical baluns back to back via their "balanced" ports, measure the loss of this combined configuration using a Vector Network Analyzer (VNA), and then divide that loss by two to get the loss of a single device (refer to this post on the Johanson website: Chip Balun, Definitions and Measurement Terminology)

But this method requires two baluns.  Suppose I only had one?  Or, if I had two baluns, how would I know if the two baluns were truly identical?

So I wondered -- what would happen if I simply connected the two ports of the Nooelec 1:9 Balun directly to my VNA?  Looking at its schematic, it looked like there might be two issues.

First, the balun's secondary winding (that is, the winding connecting to the balun's high-impedance port) of the Nooelec's balun transformer has a center-tap, and this center-tap is tied to ground.  If I connect a VNA port directly to the balun's 450-ohm (balanced) output, the VNA port's ground would short out half of the secondary windings.  Not a good idea.

I could avoid shorting out half the secondary if I cut the trace connecting the center-tap to ground, as shown, below (and that's what I did).

But there is still a second potential problem -- I would still be connecting a balun's "balanced" output (albeit with transformer center-tap now floating) to an unbalanced port of the VNA, and in doing so the balun's balanced port (balanced with respect to ground) would be forced to be unbalanced by the direct connection of one side of that port to ground..

But why not give it a try (after first floating the center-tap), and see what happens?

The results were not good.  If I placed my hand on either of the two coax cables connecting the VNA to the Nooelec balun, the s-parameter measurements would change!  (Important note:  VNA measurements should not change if you touch the coax cables connecting a Device-Under-Test to the VNA.)

The s-parameter plots, below, demonstrate this problem (note the inset photo of the test setup).

First, here are the balun s-parameters without hand touching coax...

...and next, with hand touching coax:

A huge difference!

Adding snap-on common-mode chokes to the coax cables reduced the common-mode coupling, but there were still some troublesome anomalies that I could not account for, and also, how would I know if the choking action was sufficient?

Clearly, connecting the Nooelec balun's ports directly to the VNA was problematic.

Maybe I could attach a transformer to the balanced-port of the balun to isolate it from the VNA, like this:

After all, this is essentially what happens when you connect two baluns back to back.

But the transformer would have its own impairments, such as loss and inductance.  Again, problematic.

About this time I was discussing the topic of balun measurements with Dick Benson, W1QG, and he mentioned that he had run a series of balun measurements back in 2018 using MATLAB and a technique he developed in which a 2-port balun is treated as a 3-port network.

What got my attention was that this technique allowed the balun's "balanced" port to always remains balanced with respect to ground.  And MATLAB allowed him to mathematically transform these results into a 2-port network from which loss and common-mode impedance could be directly computed.

This sounded ideal!

So...what is Dick's balun-characterization method and how do we use it?

First, Dick characterizes the 2-port balun as a 3-port network.  Let's look at how (and why) this is done:

Characterizing a Balun as a 3-port Network:

We normally think of a balun as being a 2-port device with one of its ports balanced with respect to ground and the other port unbalanced (e.g. one side of the port tied to ground).  Dick's technique treats the balun as a 3-port device.  That is, in addition to a balun's "unbalanced" port, each of the two terminals of the balun's "balanced" port is now considered to be its own port.

As an example, let's consider the "Nooelec 1:9 Balun."  The left-hand port is the unbalanced (with respect to ground) port, and the right-hand port is the balanced port:

And here is its 3-port representation:

How does this 3-port model help us?

To calculate power-loss and common-mode impedance, we need to first measure the s-parameters of this 3-port network.  If we can ensure that ports 2 and 3 always see the same impedance (with respect to ground) during the measurement process, then we will have ensured that the balanced port has remained in balance.

The method of measuring the s-parameters of this 3-port balun with a 2-port VNA is one that ensures that all 3 ports will always see an impedance of 50 ohms to ground, irrespective of the 2-port measurement being made.

The drawing below is an example of the Balun's three ports always seeing 50 ohms during a VNA measurement:

If all three ports always see an impedance of 50 ohms to ground (during VNA measurements), then the balun's "balanced" port is always balanced with respect to ground.
(Note -- this technique does not guarantee that a balance must always exist, but it should significantly reduce balanced-port imbalances.  After all, if the balun is mounted on a PCB and there is more ground-plane near one of the balanced terminals compared to the other terminal, an imbalance will exist due to differences in capacitive coupling to ground.  The 50 ohm terminations should help "level out" such imbalances, if these stray couplings are high impedances, but they cannot completely correct for poor balun design.)
 We need to represent the 3-port network with its s-parameter matrix, which looks like:

How do we measure these nine s-parameters with a 2-port VNA?  Fortunately, there is a common procedure for doing this.

First, we measure the 2-port s-parameters (S11, S21, S12, S22)  between port 1 and port 2 of the 3-port network (for the Nooelec balun, this would be between the balun's unbalanced-port port and the port representing the first (of two) "balanced" terminal).  Port 3 is terminated with 50 ohms.

The second step measures the s-parameters between port 1 and port 3 of the 3-port network (i.e. between the balun's unbalanced-port and the port representing the second "balanced" terminal).  Port 2 is terminated with 50 ohms.

And finally, the third step measures the s-parameters between port 2 and port 3 of the 3-port network (i.e. between the two terminals of the Nooelec balun's balanced output).  Port 1 is terminated with 50 ohms.

Like this:

At the end of these three steps we will have collected 12 s-parameters (i.e. S11, S21, S12, and S22 for each of the three port-pairs).  We then use 9 of these measurements to populate the 3-by-3 s-parameter matrix.

Below is a photo showing the Nooelec Balun as a 3-port Network (with two SMA connectors added to the "balanced" port).  It is connected to a 2-port VNA via the two coax cables.  Note that the 3rd port of the balun is terminated with 50 ohms.

The following code block illustrates how Dick's code fills the 3x3 matrix (named 'balun_sp') from the two-port measurements

Note that SAB represents the 2-port measurements between ports 1 and 2 of the 3-port network.  SAC represents the 2-port measurements between ports 1 and 3 of the 3-port network.  And SBC represents the 2-port measurements between ports 2 and 3 of the 3-port network.

Determining Balun Loss

Now that we've populated the 3-port network's s-parameter matrix, what is the next step?  Loss depends upon load impedance, but an impedance (i.e. R + jX), conceptually, is a one-port network.  How do I connect a 1-port load to the  2 output ports of my 3-port network?

We need to somehow convert the two output ports back to one single port.  Then I can connect the load to  this port.  But this conversion must be done in such a way as to not throw the balun's "balanced" port into imbalance.

Dick's technique relies on a mathematical model of an "ideal" (no loss, no delay, no inductance) 1:1 transformer to convert the balun back to a 2-port network without losing the balance of the balanced port.

Conceptually, the balun network would look like the transformer-coupled balun I mentioned earlier in this post.  But this transformer isn't physical, it is mathematical and avoids the impairments a physical transformer would introduce:

To connect this "mathematical" transformer to the 3-port balun, Dick implements the transformer as a 3-port entity within MATLAB.  Being a 3-port network, two of the transformer's ports are then "virtually" connected to the two output ports of the 3-port balun.  And the third port of the transformer becomes an "unbalanced" port that connects to load.

Here is the 3-port network representation of the "ideal" 1:1 transformer:

The diagram below shows the connection of the balun's two output ports to the two input ports of the transformer (this connection is made mathematically using MATLAB's cascadesparams function).  The result is a 2-port network with one unbalanced input port and one unbalanced output port.

With this new network and using MATLAB's powergain function (and specifying a load impedance for the function's 'zl' input variable), we can calculate Gp (Operating Power Gain) given the load impedance.

Please note that this transformer is a mathematically "perfect" MATLAB model of a 1:1 transformer.  It has neither loss nor delay.  And given its 1:1 turns-ratio, its 3-port S-parameter matrix is:

Note, we can generalize the 3-port S-parameter matrix for an ideal transformer with an N:1 turns ratio as:

If you substitute '1' for N in the above equations, you'll get the numbers shown above in the 3x3 s-parameter matrix for the 1:1 transformer.  (More on the derivation of these equations later in this post).

Here's the MATLAB code to create the 3x3 s-parameter matrix for the transformer (given the turns-ratio 'N'), and cascade the two 3-port networks (the cascading is done with the last line of code).

Power loss is calculated as "Operating Power Gain" (Gp).  Loss depends upon load, and the MATLAB code calculates power loss for a set of given load resistance values (defined elsewhere in the code).

Note that a single line of MATLAB code performs the power-loss calculation.

OK, so we can treat the 2-port balun as a 3-port network, make a series of 2-port s-parameter measurements with a VNA (while keeping the balun's "balanced" port always balanced with respect to ground), and then use MATLAB to calculate balun's power-loss for a given load.

But what about measuring Common-Mode impedance?

Measuring Common-Mode Impedance:

To determine a balun's Common-Mode impedance, one technique is to measure the balun's reflection-coefficient when configured in "common-mode" (in which the input is shorted together and floated from ground, and the output is shorted together and tied to ground:

In other words, we are treating the balun as a single-port impedance (R + jX) tied across port 1 of the VNA and measuring its S11.  From S11 we can calculate its impedance.

But our balun is now a 3-port network.  How should we short the output ports?

 Dick mathematically creates a 3-port "T" connection and then cascades it with the balun's 3 port network, using MATLAB's "cascadesparams" function.  This "T" shorts the two balun output ports and attaches them to a third port, so that the resulting "cascaded" network is again a 2-port network.

The 3-port T's s-parameter matrix is:

To calculate common-mode impedance, rather than drive a "floating" Port 1, Dick instead shorts Port 1 to ground and measures the reflection coefficient at Port 2 using MATLAB's gammaout function. (Note that Port 1 is shorted to ground by setting the 'zs' input variable of the gammaout function to 0).

The resulting S22 measurement is the balun's common-mode impedance.

Here's how it looks:

Note that, functionally, this is the same as shorting the balun's output to ground, floating the input port, and measuring S11 from the floating input-port to ground.

Measurement Results:

I soldered two SMA jacks to the back of the Nooelec 1:9 Balun to turn it into a 3-port network (keeping its transformer's center-tap grounded) and then ran the series of s-parameter measurements.

Dick's MATLAB code takes these s-parameter measurements and plots a number of results versus frequency, as shown below:

Balun Power Loss

This is a plot of Gp (Operating Power Gain), but here it is plotted as a loss.

Note that I've calculated (and plotted) both Forward and Reverse power loss.  The turns-ratio of the ideal transformer used in the MATLAB code is 1:1.

Given the balun's usual application of being a receive balun with an antenna connected to its 450-ohm balanced port and a receiver (with a 50-ohm input?) connected to the balun's SMA, the curve-of-interest would be the cyan plot, not the yellow plot.

Common-mode Impedance:

Below is a plot of the balun's common-mode impedance.  Note that the calculation that  the common-mode impedance be measured at the network's output port.  So this is a measure of the network's S22.

Note how low the common-mode impedance is.  If you need common-mode rejection, add proper common-mode chokes.


Below are plots of the balun's s-parameters (derived from the 3-port measurements).

Note that the loads to the input and output ports of the balun are both 50 ohms (at the output port this is because the "ideal transformer" has a turns ratio of 1:1).

The balun's S22 is around 450 ohms because the 50 ohms at the balun's input is multiplied by 9 (at least theoretically) by the balun's transformer.

And the balun's S11 is around 5.6 ohms because the VNA's 50 ohms connected to the balun's output port is divided by 9 by the balun's transformer.

S12 and S21 losses are in the 4.6 dB range because of the reflected-power at the input and output ports due to S11 and S22 not being anywhere near 50 ohms (power reflected back from the input port, for example, also manifests as a worse S21).

S11 of the balun's 50 ohm input port with a 450 ohm load at the balun's output:

The MATLAB code allows me to terminate the balun with a perfect 450 ohm load, resulting in the S11 plot, below.

You can see that the network's input impedance with a 450 ohm load is close, but not quite 50 ohms.

S11 of the balun's 50 ohm input port versus other load impedances:

This plot shows S11 for output loads from 350 to 600 ohms, in steps of 50 ohms.  Interestingly, the input's match to 50 ohms looks best when the load is 550 ohms (i.e. the red line), rather than 450 ohms.

Balun Power Loss versus load impedance:

And here is balun power loss for different output loads (from 350 to 600 ohms, in steps of 50 ohms), as well as a plot of power loss when the output port is driven and the SMA port sees 50 ohms.

Interestingly, as load-impedance drops, so does power-loss.

S-parameters with 3:1 Ideal Transformer:

The mathematical model of an ideal transformer used for the above plots has a turns-ratio of 1:1, but there's no reason why we couldn't use a "mathematically perfect" transformer with a different turns ratio to get a better match between the balun's output port and the VNA's 50 ohm impedance.

If a 3:1 "ideal" transformer is used in lieu of the 1:1 "ideal" transformer, then both S11 and S22 should be significantly close to 50 ohms.  Also, S12 and S21 should show less loss, because less power is being reflected back due to port mismatch.

The plot below compares the parameters of the balun with the 'ideal" 1:1 transformer (yellow traces) versus the balun with an "ideal" 3:1 transformer (cyan traces).  You can see that the 3:1 transformer brings the balun's input and output port impedances much closer to 50 ohms.

Balun Power Loss with 3:1 Ideal Transformer:

Here's a plot of power-loss with the 3:1 transformer replacing the 1:1 transformer.  And if you compare the plot below with the earlier plot made with the 1:1 transformer, the losses are essentially identical.

Comparing 3-Port Measurements with Back-to-back Balun Measurements:

Nooelec's measurements of loss were made by connecting the 450-ohm ports of two-baluns together and then attaching the VNA ports to the baluns' SMA connectors:

I thought I'd repeat this measurement, but I would calculate Gp rather than Insertion Loss.

Because input and output impedances of this back-to-back balun configuration should be around 50 ohms, it made sense to me to compare it to the 3-port balun that was cascaded with the "ideal" 3:1 transformer, which also has input and output impedances around 50 ohms.

So I purchased a second Nooelec balun and connected the two baluns as shown, above.  The results are below:

S-Parameter Comparison, Balun as a 3-port Network versus Two Baluns Back-to-back:

First, a comparison of S-parameters.  Note that the insertion loss (S21 and S12) of the back-to-back configuration is almost exactly twice the insertion loss of the single balun.

S11 and S22 are also worse for the back-to-back baluns.

Power Loss Comparison, Balun as a 3-port Network versus Two Baluns Back-to-back:

Interestingly, when I compare the power-loss of the two configurations (after first dividing the Gp loss of the two back-to-back baluns by 2 to approximate the loss of a single balun), the back-to-back baluns look better than the single balun.

I can explain why mathematically...S21 and S12 are essentially equivalent for the two configurations (if we first divide S21 and S12 of the back-to-back configuration by 2), but the worse S11 and S22 of the back-to-back configuration result in the calculated power-loss being less.  (See the formula for Gp, below).

Anyway -- that's the mathematical explanation.  Not very satisfying.

Other topics...

1.  What is Gp (Operating Power Gain) and Why do I Prefer it to Insertion Loss?

I am interested in the power dissipated within a balun.  If there is too much power, the balun breaks.

Gp is the power delivered by a network to its load minus the power entering into the input of the network. (Note that the power entering into the network's input is the power arriving at the input port minus the power reflected back from this network's input.  Whatever power isn't reflected by the input is power entering into the balun).

If Gp is positive, there is gain.  And if Gp is negative, there is loss.

If there is a loss, this loss represents the power lost within the network.  It is this power loss that heats up the network.

Here is the equation for Gp per the Mathworks website (to which I've added a few notes):

On the other hand, Insertion Loss includes input power that is reflected back by the network's input (due to input mismatch).  It does not enter the network, but it is included in the Insertion Loss quantity.  And because this reflected power is not dissipated within the balun, I do not want to include it in the balun's loss calculation.

2.  Deriving the 3-port Ideal N:1 Transformer S-parameter Matrix:

Dick used MATLAB's Symbolic Math to create the equations for the 3-port s-parameter matrices for the ideal transformer and for the ideal T connection.  But Dick is a much more experienced MATLAB user than am I.  So I've instead derived these s-parameters the old fashioned way, with quill and parchment.

As a reminder, a 3-port network has 9 s-parameters, and they are arranged in a 3x3 matrix as follows:

The diagram below recaps the port assignments of the ideal N:1 transformer.  Note that port 3 represents the transformer's "unbalanced" port.

Next are equations for the elements of the 3x3 s-parameter matrix:

And finally, here are my derivations of the above equations.

First, S11 and S22:

Next, S33:

Then, S21, S12, S31, and S32:

Note:  When Zload = Zsource = Zo, S21 = 2*(V2/Vg)  (See page 17 of:
Ditto for the other Transmission s-parameters.

And finally, S13 and S23:

3.  Deriving the 3-port T's S-parameter Matrix:

Below are my derivations of the s-parameters for the 3-port ideal "T" connection.

First, a recap of the port assignments.

Next are values for the elements of the 3x3 s-parameter matrix:

And finally, here are my derivations of the above equations.

First, S11, S22, and S33:

Followed by the rest of the S-parameters,  S31, S13, S32, S23, S21, and S12:

4.  MATLAB Script to Generate 3-port Transformer S-parameter Equations:

As I mentioned above, Dick used MATLAB's Symbolic Math to generate the 3-port Transformer's s-parameter equations.  Below is his MATLAB script.

(I've added a note in the script identifying the other MATLAB resources required to run it.)

% 3 Port Transformer Symbolic Analysis
% Dick Benson June 2020
% Note that this script requires MATLAB's Symbolic Math
% Toolbox and RF Toolbox as well as Dick's "Function Based
% Symbolic Circuit Analysis" package of scripts (etc.),
% available for download from the Mathworks MATLAB Central
% File Exchange site.
% - Jeff, k6jca
clear; close; clc;

syms N Z Zo real

    % S11 = (Z-Zo)/(Z+Zo);   Ports 2 and 3 terminated in Zo
    % Port 1  Z =  (Zo*N^2)  +  Zo
     Z = Zo*(N^2) + Zo;
     S11 =  simplify((Z-Zo)/(Z+Zo));
     S22 =  S11; % by inspection

     % Port 3  Z = (Zo+Zo)/(N^2)
     Z   =  2*Zo/(N^2);
     S33 =  simplify((Z-Zo)/(Z+Zo));

     % S21 = Power delivered to Port 2 from Port 1
     %       with Port 3 terminated in Zo.
     %   Equivalent a series resistor ----Zo*N^2----
     %   between ports 1 and 2.
     % Y parameters for a series-R
     Y_series_R = [1 -1; -1 1]/(Zo*N^2);
     S = y2s_symbolic(Y_series_R,Zo);
     S21 = S(2,1);
     S12 = S21;

     % S13 = power delivered to port 1 from port 3
     %       with port 2 terminated in Zo
     % This is equivalent to having a 1:N transformer
     % with a series Zo on the secondary side.
     abcd_trans = [1/N 0; 0 N];
     abcd_series_Zo = [1 Zo; 0 1];
     abcd_cascade = abcd_trans*abcd_series_Zo;
     S_cascade = simplify(abcd2s_symbolic(abcd_cascade,Zo));
     S13 = S_cascade(1,2);
     S31 = S13;

     % S23 is simply S13 with 180 degree phase shift
     S23 = -S13;
     S32 = S23;

     S_Result =   [S11  S12  S13;
                   S21  S22  S23;
                   S31  S32  S33]
S_Result =
[   N^2/(N^2 + 2),      2/(N^2 + 2),      (2*N)/(N^2 + 2)]
[     2/(N^2 + 2),    N^2/(N^2 + 2),     -(2*N)/(N^2 + 2)]
[ (2*N)/(N^2 + 2), -(2*N)/(N^2 + 2), -(N^2 - 2)/(N^2 + 2)]

5.  A Note regarding Common-Mode Z derived from S11 Measurements:

If Common-Mode Z is high, beware that an S11 measurement might be inaccurate due to resonance with the test fixture.

I have personally found that there is about 2 pF of capacitance, to ground, at the port where I measure S11.  This capacitance can form a parallel-resonant circuit with the choke's inductance and lower the "peak-impedance" frequency.

If it appears that you might have a resonance (i.e. sharp peak, for example, and phase passes through zero at the peak's frequency), I'd recommend using G3TXQ's method of deriving CM Z using S21 (in lieu of S11).

For more on this techniques, see this blog post:



I have uploaded Dick's MATLAB 3-port Balun Analysis code and examples to my Github site.  You can download them from:

The MATLAB code written by Dick Benson requires, of course, MATLAB.

In addition, it requires the associated MATLAB RF Toolbox and also the "S-Parameter Utilities" that Dick has uploaded to MATLAB Central: (see below).  You can download the "S-Parameter Utilities" package via the link,
  • S-Parameter Utilities (Very useful!  A number of these utilities, such as s1p_viewer, are used by Dick's other MATLAB scripts).
Dick has also written a number of other packages available on the Mathworks MATLAB Central File Exchange site (here).  Of particular interest are the following:
The latter two uploads of Dick's require MATLAB's "Instrument Control Toolbox" in addition to the "RF Toolbox".

Standard Caveat:

I might have made a mistake in my designs, schematics, equations, models, etc.  If anything looks confusing or wrong to you, please feel free to leave a comment or send me an email.

Also, I will note:

This design and any associated information is distributed in the hope that it will be useful, but WITHOUT ANY WARRANTY; without an implied warranty of MERCHANTABILITY or FITNESS FOR A PARTICULAR PURPOSE.

Saturday, May 23, 2020

ME-165/G Standing Wave Ratio - Power Meter

The ME-165/G is a military SWR and Power meter, designed for the HF bands (1.5 - 30 MHz), that includes an internal 600 watt dummy load.

The image, below, shows the ME-165/G as part of the AN/GRC-26D shelter-mounted radio teletype station:

The unit provides a convenient way to switch the dummy load in and out of the transmission line, plus, if tuning an antenna tuner in SWR mode, its SWR circuitry allows the transmitter to always see a 50 ohm load, irrespective of the actual load at the unit's Output port.  So you don't need to worry about destroying your finals if you make a mistake while tuning your antenna tuner.

There are four modes of operation (per the four positions of the front panel's rotary switch).  The table below describes how the ports connect and the meter function for each of these four modes:

The following illustration shows this same information as a "functional diagram" (i.e. part block diagram, part schematic).

Antenna Tuning Procedure:

When tuning an antenna tuner, the ME-165/G should first be placed into "ADJUST" mode and the "ADJUST" potentiometer rotated for a full-scale meter reading while transmitting a CW signal.

Then turn the rotary switch to its "SWR" position and adjust an external antenna tuner (connected between the ME165/G's Output connector and the antenna) to give a minimum reading on the SWR meter (for a correctly tuned tuner, the meter's needle should end up in the green-region at the left-hand side of the scale).

Here is a closeup of the Power and SWR meter scales:

Note that the SWR scale is NOT accurate above about 2:1.  (I'll discuss this in more detail later in this blog post).


The schematic, from the Army's Technical Manual.  I've corrected a couple of errors (my corrections are in red):

Also, note that C8 (the capacitor at the Input connector) is listed in the schematic as 40 pF.  In the two ME-165/G units that I have, its capacitance is actually 39 pF.

Schematic Notes:

1.  The 1200 ohm, 25 watt resistor, R15 in the schematic, in series with the SWR Bridge circuit reduces the power delivered to the SWR bridge circuit (therefore, the bridge can use 1/2 watt resistors).

2.  SWR Detection is via a Wheatstone bridge.  The bridge is balanced when the load at the ME-165/G's Output port is 51 ohms.

3.  C6 puts the ADJUST pot wiper at RF Ground, thus R22 (1200 ohm) is essentially in parallel with the  lower-right-hand side side of the bridge (via C5) -- i.e. this resistance is in parallel with whatever load is attached to the unit's Output port.

Thus, an equivalent resistance (R20, 1200 ohms) must be connected in parallel with the lower-left-hand arm of the bridge to ensure that the bridge is balanced when the external antenna impedance connected at the Output side of the bridge is 51 ohms, resistive.

ME-165/G Performance:

I own three ME-165/G's.  Let's look at their performance.

First, the unit manufactured by Radalab, Inc:

The (first) Radalab, Inc., ME-165/G (circa 1970's):

I picked up this unit many years ago.  The unit was in good shape, but a previous owner had replaced the original N connectors with SO-239 connectors.

Inside, the majority of the components are wired point-to-point using solder posts:

Below is a photo showing the wiring from one side of the Dummy Load to the rotary switch and from the other side of the Dummy Load to ground.  Note that the load's ground wire goes to C8's ground terminal.  This terminal is then grounded to the front panel via a separate wire.

The other wire from the load (to the rotary switch) is bound tightly with the ground wire from the load, using three cable ties.

The 50 ohm, 600 watt load consists of twelve Dale 600 ohm, 50 watt resistors in parallel:


Measuring the unit's S-Parameters in dummy-load mode with an HP 8753C Network Analyzer:

Below is the capture of S11 when the ME-165/G's rotary switch is set to POWER (i.e. the Input port is connected to the 50 ohm Dummy Load).  You can see that the SWR in the HF band (to 30 MHz) looks very good (1.1:1 at 30 MHz).  Not so good at 6 meters, but this load is not spec'd to that frequency.

When the front-panel rotary switch is in the OPERATE position, the dummy load is disconnected from the Input connector and the Input is connected directly to the Output connector.

How does the Radalab ME-165/G affect performance when it is in its OPERATE mode?  Again, let's look at the s-parameter measurements...

As you can see in the plot below, there is some insertion loss that worsens with frequency, but this loss is only about 0.09 dB, worst case, at 30 MHz.

And the SWR of the "ideal" 50 ohm external load, rather than being 1:1, is now changed by the Radalab ME-165/G to be 1.2:1 (at 30 MHz).

So, some minor adverse effects, but overall, not bad! 

Improving the performance of this (first) Radalab ME-165/G in OPERATE mode:

I had noticed a difference between the Oneida's output wiring and the wiring of the Radalab unit, and I wondered if this difference accounted for the slightly worse Radalab Insertion Loss when in OPERATE mode.

The wiring from the rotary switch to the Oneida unit's output connector had been routed next to the front panel.  But the same wire in the Radalab unit was routed high in the air, as shown below:

Would moving this output wire to be closer to the front panel make a difference?  Here's a photo of the new routing:

And below is the measured s-parameters for this new routing.  Note that Insertion loss has decreased from 0.09 dB to about 0.07 dB.  Note much of a change, but it's in the right direction.

And the image below shows that the SWR of the "ideal" 50 ohm load is now 1.1:1 (from 1.2:1).  Again, a slight improvement, but an improvement never-the-less.

Oneida Electronics Inc ME-165/G (circa 1963):

My second ME-165/G was manufactured by Oneida Electronics Inc, around 1963 (per the suffix on the Order Number listed on the front panel tag).

This unit still has the stock N-connectors on the front panel (note that in the image, below, there are BNC adapters attached).

Here's a look at the inside terminal-board used for wiring the components.  You can see that it is similar to the later Radalab's board (shown above):

But there is one noticeable difference between the Radalab board and the Oneida board, which is the use of clips to hold in CR1 and CR2, rather than soldering them to posts, as shown in the two photos, below:

CR1 (1N69A)

CR2 (1N277)

I imagine clips allowed easy replacement of the original 1N69A diodes in case they blew out.  Note that CR2 (a 1N277 diode) has its leads wrapped around the clip's posts.  This diode probably replaced a bad 1N69A diode.

The image below shows Oneida's wiring from the 600 watt dummy load to the rotary switch and ground.  Note the difference between this wiring and the wiring in my Radalab unit (shown earlier in this post).  The wires below are not routed together in parallel, and the dummy load's ground wire goes directly to a ground terminal on the front panel, rather than first routing to C8's ground.

Also, the dummy load's resistors are not Dale resistors; instead they are TRU-OHM 600 ohm non-inductive resistors:

Improving the performance of the Oneida ME-165/G:

When I first measured the SWR of the dummy load, I noticed that it rose to 1.5:1 at 30 MHz.  So I tried to improve its SWR by changing the dummy load's wiring to look exactly like the wiring in my Radalab ME-165/G.

But with this modification the SWR rose to 1.75:1 at 30 MHz (see below).  Yikes -- my attempt to make the Oneida's wiring match the Radalab wiring moved SWR in the wrong direction!

OK -- mimicking the Radalab's wiring was not going to work.  Playing around with the separation of the dummy load's two wires, I discovered that (a) separating the two wires far apart, and (b) keeping the original ground wire to the dummy load, in addition to the new (red) ground wire I'd added, improved the SWR.

The image below shows the new dummy-load wiring.  The red wires are 14 AWG THHN stranded wires (insulation rated to 600V).  Note their separation! You can see the dummy load's grounding red wire goes to C8, just like the wiring in the Radalab unit.  But you can also see that the original ground wire is still connected to the dummy load (and now routed a bit closer to the upright mounting plate).

And here is the new SWR measurement.  Now it is 1.2:1 at 30 MHz.

(Perhaps the difference in wiring is due to a difference in impedance between the Oneida's 600 ohm TRU-OHM resistors and the Dale 600 ohm resistors in the Radalab unit?)

In OPERATE mode the Oneida unit's measurements look very good.  Here's S21.  Insertion loss is only about 0.04 dB at 30 MHz.

And there is little impact on the SWR of an external 50 ohm load.  As you can see, below, the SWR at 30 MHz for the external 50 ohm load is 1.09:1.

Second Radalab, Inc., ME-165/G (S/N 47C):

The first Radalab unit I discussed (above) is serial number 7C.  This second ME-165/G is serial number 47C.

Interestingly, its component-mounting board no longer has the clips for the diode leads.  Instead, all components are attached with soldering posts:

(This difference could be due to a later unit upgrade, or it might have been a change during the manufacturing run).

Otherwise, the two Radalab units look very similar. 


Again, made with my HP 8753C Vector Network Analyzer.

Below is the S11 plot with the Function Switch set to POWER.  Note that the SWR at 30 MHz is about 1.5:1.  Not as good as I would like it to be.

Below are the s-parameters with the Function Switch set to OPERATE:

SWR (of an external 50 ohm load) is transformed to about 1.3:1 (from the load's original 1:1) at 30 MHz.

And Insertion Loss is about 0.1 dB. 

Improving the performance of this second Radalab ME-165/G:

The image below shows the original wiring of this ME-165/G:

If I routed the dummy-load wires together (using tie-wraps), SWR in POWER mode improved.

The image below shows the new routing of the two dummy-load wires:

(Note, I had to trim off a small amount of the wire going to the big black cap because it was just a too long, as you can see in the photo, below.)

With this modification, the dummy-load's SWR went from 1.5:1 to less than 1.1:1 (see below).

I had noted that the wire to the output jack was up in the air.  So I moved it to route along the inside of the front panel. 

I don't know the voltage rating of this wire's insulation, and I was a concerned that it now ran against the grounded metal of the front panel, so to prevent the possibility of arcing I added a bit of Kapton tape between it and the panel for extra voltage insulation (and I added a second piece of tape to hold the wire next to the panel), as shown below:

This modification improved OPERATE mode's Insertion Loss (from 0.1 dB to about 0.05 dB) and the "through" SWR of an external 50 ohm load (now 1.1:1 from the unmodified version's 1.3:1).

An SWR Meter that does not measure SWR:

While using the ME-165/G,  I discovered that its SWR readings can be very inaccurate'

Here's a look at the SWR scale on the ME-165/G meter.

As I mentioned, the SWR reading can sometimes by quite inaccurate.  For example, what should be the SWR when the load is a short?  Of course, it should be infinite (meter needle at full scale).  But that's not what the ME-165/G shows:

So my Radalab unit shows that the SWR of a short-circuit is somewhere between 3:1 and 4:1.  And if I repeat the test on my Oneida unit, the SWR of a short measures slightly less than 3:1 (the difference between the two is probably due to drift of component values over time).

In other words, both of my ME165/G SWR meters show an SWR of around 3:1 for a short circuit.  Neither unit shows the correct SWR of infinity (meter needle at full scale).

Despite this gross SWR inaccuracy for a short-circuit load, the SWR meter's accuracy seems to improve considerably below an SWR of about 2:1.  Therefore, as long as the goal is to tune the antenna for minimum SWR, rather than measure its SWR value, the ME-165/G does the job quite well.

But I still wanted to know -- why was the SWR meter so inaccurate for a short-circuit load?

SPICE Simulations:

I decided to do some SPICE simulations to get a better understanding of what to expect from the ME-165/G SWR detector.

The ME-165/G's SWR measurement circuit is based upon a simple Wheatstone Bridge, with the unknown load to be measured represented by the lower right-hand arm of the bridge, as shown, below:

In an ideal Wheatstone Bridge we can take the difference between Va and Vb, then divide by Va, and then take the magnitude of this value, we can create a set of numbers that we can equate to SWR values, as shown in the table, below:

Note that the quantity |(Va - Vb)/Va| is equivalent to the magnitude of the load's Gamma:

|Γ|    SWR
    0     1.00 
 0.1     1.22
 0.2     1.50
 0.3     1.86
 0.4     2.33
 0.5     3.00
 0.6     4.00
 0.7     5.67
 0.8     9.00
 0.9   10.00
 1.0   infinite

In other words, if we could measure Va and Vb with high impedance measuring circuits (so that there are no unwanted currents through either arm of the bridge that might alter the bridge's balance) and then perform the math, we'd get a number equal to the magnitude of the load's Gamma, and thus translatable to its SWR.

Sounds straightforward, but note...the equation requires a division by Va.  Is there an easy way to accomplish this division with simple circuitry?

If we could adjust our voltages so that Va equals 1 (while keeping the ratio of Va to Vb constant), then we can skip the division step, because we would be dividing by 1.

In the bridge circuit above, for example, maybe we would have a switch that we would first set to an "Adjust" position, connecting a high-impedance meter to Va and letting us scale its gain (via a potentiometer) until the meter's needle is at Full Scale, i.e. so that Va now equals 1.

And then we would flip the switch to measure |Va - Vb|, using the same gain-adjusted high-impedance meter, to give us a direct reading of  Gamma thus SWR (e.g. a meter reading of 1/2 Full Scale would equal a Gamma of 0.5, or an SWR of 3:1).

But we can see from the SWR scale on the ME-165/G meter, and from our example measuring the SWR of a 0 ohm load, that the ME-165/G is doing something very different -- something that affects the accuracy of its SWR readings.

The problem is that, for the equation Vswr = |(Va-Vb)/Va| to give accurate results, Va must be measured while Rload is connected to the Wheatstone Bridge. This is because, given the ME-165's circuitry to limit the power to the Wheatstone Bridge (i.e. the series 1200 ohm, 25 watt resistor ), any change in Rload will affect the voltage at node Vc at the top of the bridge (because a change in Rload will change the current through that arm of the bridge, and thus it changes the current (and subsequent voltage drop) through this series 1200 ohm resistor feeding the bridge).

Because we are adjusting Va to be 1 (to avoid a mathematical division), the value of Va that was set during the "Adjust" step should (ideally) be the same as the value of Va used during the SWR measurement step.

But in the ME-165/G, these two Va's are not the same.  The Va of the Adjust step is measured without Rload connected to the bridge, while Va of the SWR measurement step is measured with Rload connected to the bridge.

So Vc will be different for these two steps, and thus Va (which equals Vc/2) will also be different.

Let Va1 be the value of Va measured during the Adjust, and "Va2" be the value of Va measured during the SWR measurement step.  Because the "Adjust" step is, essentially, determining the value of Va that we will use to normalize the quantity (Va2 - Vb), the original equation  |(Va - Vb)/Va| becomes:

Vswr = | (Va2 - Vb) / Va1 |

I can simulate the result in LTSpice by adding another arm to represent the "unloaded" Va (i.e. Va1).  Below is the model, and I've annotated it with the simulation results of this new equation.

We can see that the measured SWR values are different for loads with the same actual SWR (e.g. 0.34 for 150 ohms versus 0.21 16.67 ohms -- both loads have an actual SWR of 3:1), and it explains why the ME165/G's measured SWR of a short is so far off from what it should be.

Let's now add the diode detector and meter circuit to the simulation and see how they affect performance.  Please note:
1.  LTSpice doesn't seem to have any Germanium diode models, so I'm using a Schottky diode (1N5817), instead.  (Note:  if replacing the original CR1 or CR2, I'd recommend using a 1N5818 or 1N5819 for their higher peak-reverse-voltage specifications.)
2.  I've adjusted the amplitude of the driving voltage source so that R23 (representing the "Adjust" potentiometer) is 0 ohms and the meter current is 1.0 mA when Rload = 1 Megohm (by setting the current equal to 1 mA for Rload = 1 Meg, I am effectively mimicking the "Adjust" step of the SWR measurement).
3.  The 1 mA meter is represented by Rmeter (58 ohms), per my measurement of the meter's resistance.  And I've increased the meter's bypass cap (C7) from 1 nF to 100 nF to knock down the RF across the meter and make it easier to determine the DC current passing through Rmeter.
4.  The frequency of the sine-wave drive is 4 MHz.
5.  Circuit parasitic elements are not included in the simulation.
I would expect the addition of the diode-detector to throw off the simulated values determined earlier (for the "ideal" Wheatstone Bridge), because the diode will conduct during part of the RF cycle, squirting current from the right arm of the Wheatstone Bridge (Vb) into the left arm (Va) and thus changing these two voltages.

Here's the new LTSpice schematic:

And below are some simulations of this new circuit...

First, verifying that the "meter" current is 1 mA when mimicking the ME-165/G in ADJUST mode, i.e. when there is no load (Rload = 1 Megohm):

Next, replacing the "open" load with a short.  Ideally, the current should remain 1 mA (representing an infinite SWR).  But as you can see, the DC current is 0.4 amps, which is quite a ways off from the 1 mA target.

Let's take a look at two loads that should each have an SWR of 3:1:

First, a Load = 150 ohms (note that the meter current is 0.32 mA):

Next, a Load = 16.67 ohms (note that the meter current is 0.21 mA):

These results are not exactly the same as the results made without the actual diode-detector in the circuit, but they are close.  (I believe the difference is due to the actual diode-detector acting as a current path between the two arms of the bridge, when in fact these two arms should be isolated from each other).

Below is a table of simulation results, simulated at 4 MHz and at 10 MHz, for different load resistances.  The third column is the actual DC current required to drive the meter's needle to the appropriate "tick" mark on the ME-165/G meter's SWR scale.  If you compare the "required" current to the "actual" (i.e. simulation) current, you can see that only some of the simulated currents come close to target values.  Only when the load's actual SWR is about 2:1 or better do we seem to get in the ballpark of the actual meter tick marks, irrespective of whether the load is greater than 50 ohms, or less than 50 ohms.

The simulated results also depend upon the type of diode used.  As I mentioned earlier, LTSpice does not seem to have a Germanium diode model, so I used a 1N5817 Schottky diode instead.

I thought I'd look at the simulation results using other LTSpice diode models.  The table below shows simulation results of two different Schottky diodes (1N5817 and BAT54), and a common 1N4148 Silicon diode.

Note the loss of resolution at low SWRs if using the 1N4148 diode.  This will result in tuning appearing to give a 1:1 SWR over a broader range of loads, which is not desired!


1.  The ME-165/G provides a 600 watt dummy-load and power-measurement meter for the HF range of 1 to 30 MHz) that can be easily switched in and out of the transmission line.

2.  It might be possible to improve either the dummy-load's SWR or the "through" insertion loss at the high end of the HF range by changing wire routing.  Use a Vector Network Analyzer (such as the NanoVNA) to accomplish this by measuring S21 and S11.

3.  The ME-165/G provides an SWR measurement mode useful for adjusting antenna tuners.  However, the meter's accuracy very much depends upon the load value.  Accuracy seems to improve as the SWR drops below 2:1.

4.  If replacing diode CR2 in the SWR circuit and you cannot find the original 1N69A (or 1N277), try using a Schottky diode such as a 1N5818 or 1N5819, rather than a generic silicon diode such as the 1N4148.  (On the other hand, a 1N4148 diode should be fine as a substitution for CR1).

I recommend the 1N5818 or 1N5819 instead of the 1N5817 I used in my simulations because these two diodes have a higher peak-reverse-voltage specification compared to the 1N5817.  Although PRV of the 1N5817 is 20 volts and the worst-case simulated peak-reverse-voltage was around 14 volts (for Vin = 250Vpp, F = 2 MHz, Rload = Open, and the bridge resistors assuming a worst case 10% variation (R19 = 56 ohms, R21 = 46 ohms, and R18 = 46 ohms)), I personally would prefer to have a bit more PRV margin.


     Technical  Manual TM 11-6625-333-15

     PA0FRI ME-165/G website

Standard Caveat:

I might have made a mistake in my designs, schematics, equations, models, etc.  If anything looks confusing or wrong to you, please feel free to leave a comment or send me an email.

Also, I will note:

This design and any associated information is distributed in the hope that it will be useful, but WITHOUT ANY WARRANTY; without an implied warranty of MERCHANTABILITY or FITNESS FOR A PARTICULAR PURPOSE.