K6JCA

New QSL Card!

2012-01-24T09:49:00.001-08:00

Finally, a new QSL card:

(My design. Printed by KB3IFH).

Quickie Pneumatic Antenna Launcher

2011-06-09T14:06:00.000-07:00

I need to get wire-antenna supports up into some tall pines at a remote location, and the slingshot that I would normally use to do this is at my brother's house. So...in its absence I thought I'd instead make a "pneumatic antenna launcher" to help me get the supports up into high tree branches.

A quick Google search revealed a number of plans for pneumatic antenna launchers, the most common using 2.5" PVC pipe. Although these designs were usually pretty fancy (using adapted sprinkler valves to trigger the launchers), I thought they might form the basis of a simpler design that I could quickly assemble. So off to Home Depot I went to pick up some 2.5" PVC and accessories.

Unfortunately, when I arrived I discovered that the local Home Depot only has Schedule 40 PVC pipe up to 2" inner-diameter, but not 2.5" pipe.

Well, why not use 2" pipe? With this diameter in mind, I searched through the bins of various PVC couplings and parts, designing the launcher in my head as I discovered what bits and pieces Home Depot had in stock.

With money dispensed, home I went, and not much later I had my launcher! Here it is:

(Click on image to enlarge)

The air-chamber and barrel are made from 2" I.D. Schedule 40 PVC pipe. Overall length is 90 inches. The barrel is 32 inches long, and the air-chamber is 52 inches long (roughly 2.5 quarts in volume).

I chose 2.5 quarts as a compromise between air-volume and length of the chamber. Other designs that I found on the internet seemed to use a volume of about 3 quarts for their air chambers, but, with 2" PVC pipe, this would require a chamber length of 60 inches, which I thought would make the overall launcher a bit too unwieldy. So I shortened it up a bit, which, for me, puts the "trigger" at a nice height when the end of the launcher is resting on the ground.

For the "trigger," rather than try adapting an expensive sprinkler valve as others had done, I went with a low-tech, low-cost ball valve which I'd seen used in the following photo of a potato launcher.

(Click on image to enlarge)

James and Devin with potato launcher (circa 1999?)

I chose a 1/2" ball-valve after I discovered, while testing various size valves at Home Depot, that it was the one that I could turn the easiest:

(Click on image to enlarge)

The 1/2" ball-valve is threaded at both ends. To connect it to both the 2" air-chamber and the barrel, I screwed into each end of the valve 1/2" (threaded) to 3/4" (female slip) adapters (with a generous amount of Teflon pipe-tape on the threads), and then I glued short lengths of 3/4" PVC pipe into the slip-joint ends of these adapters. In turn the other ends of these short lengths of 3/4" pipe are glued into 3/4" (slip) to 2" adapters. The barrel and the air-chamber connect to these 2" adapters via 2" slip couplings (again, glued).

Note that the threaded couplings allow the launcher to be disassembled for easier transport. And, should I ever decide to change to a fancier trigger mechanism, they would allow me to easily swap out the original ball-valve trigger for something different.

To fill the air-chamber I used a Presta valve from an old bicycle inner-tube that I had lying around. It's threaded and has a nut, which eases its installation.

(Click on image to enlarge)

A Schrader valve would have been preferred, as Presta valves are a bit fragile, but the Presta valve was what I had on hand.

To ensure a good seal between the valve and the air-chamber pipe, I cut out two pieces of the bicycle inner-tube rubber, each piece roughly a circle 1" in diameter. Into the center of each piece of rubber I cut a small hole slightly smaller than the diameter of the Presta valve. I pressed these each over the valve and worked them, one at a time, down the stem to the end that would be within the air-chamber pipe.

I drilled a small hole in the pipe just past the point where the end-cap would stop (do NOT attach the end-cap yet before you install the valve!), and then I inserted the valve into this hole. With its nut tightened down, the rubber "gaskets" I'd made provided a good seal against the inside of the air-chamber.

After I'd installed the valve, the air-chamber was capped off with a 2" PVC cap, glued in place.

Because the pipe is only 2" in diameter, I couldn't use normal size tennis balls. A visit to Jon, K6JEK, and his wife revealed exactly what I needed. Their dog Buster likes to chase 2" tennis balls.

(Click on image to enlarge)

I tested one of these tennis balls in the launcher, and it worked great! Buster was too attached to his tennis ball for me to try to take it (and his others were chewed beyond recognition), so it was off to the local Petco (pet supply) store to search for more 2" tennis balls!

(Click on image to enlarge)

The yellow balls are a bit softer than the blue/pink ball, and they are "squeaky" toys. I drilled a couple of holes in one so that I could insert a tie-wrap to use as an attachment loop. Then, at the other end, I cut a thin slit with an X-acto knife so that I could insert pennies to add weight. Per another website, the ball should weigh between 4 and 5 ounces (as the best tradeoff of height, safety, and the ability to pull the line down over tree branches and foliage). Getting it up to 5 ounces pretty much fills up a 2" tennis ball with pennies! (Each penny is roughly 0.1 ounces).

Here's a finished tennis ball, with tie-wrap attachment loop:

(Click on image to enlarge)

Using a bicycle tire-pump, I've tested my chamber up to about 80 psi and it seemed to hold its pressure fine (at least for the time it took me to insert a ball and launch it). The 2" pipe itself is rated to 280 psi (and the 3/4" pipe to 480 psi), but the ball-valve is only rated to 150 psi. I'd recommend keeping the max pressure well below this point, though.

A small paint bucket can be used to hold the line and keep it from becoming entangled in ground debris (e.g. twigs and leaves). Tie one end of the line to the bucket handle!

(Click on image to enlarge)

Results:

Shooting the weighted 5 oz. tennis ball straight up into the air resulted in the following heights:

20 psi: 15 feet
40 psi: 35 feet
60 psi: 65 feet

(Note: I only tested once at each psi level. Heights are approximate, based on a rough measure of how much line played out).

While erecting my 80 meter full-wave loop, I discovered that I needed the ball to be heavy so that, if it were in an environment with many branches, it had a better chance of pulling down the line attached to it. I had started with a 4 oz. tennis-ball load, but finally decided I was better off with the ball loaded with as many pennies as I could fit into it. The result is a ball which weighs about 5.5 oz.

Even at this weight, sometimes the ball wouldn't drop all the way to the ground, and I would have to "finesse" it down by wiggling the line or trying other tricks. And sometimes I just had to pull the ball back and start over again. Perhaps a more "slippery" line might have helped the ball descend, but in the end I was able to get all of the supports up and the loop raised without either having the ball become permanently stuck in a tree, or my having to run to the store to purchase yet one more thing.

Ready, aim...

Notes:

1. Mechanically, the weakest point is the smaller-diameter pipes and adapters that make up the trigger mechanism: this is where you'll see the launcher bending. To protect these parts when transporting or storing the launcher, I'd recommend unscrewing the barrel from the ball-valve, and not unscrewing the air-chamber. Keep the air-chamber screwed into the ball-valve, because it's important to maintain a good air-tight seal at the threads to prevent pressure loss.

2. More height-per-psi might be achievable with a better (faster) trigger mechanism (e.g. adapted sprinkler valve), but I'm satisfied with my results -- they work for my application, and the design is very simple and easy to construct. Also, because the tennis ball is narrower than 2", air can escape around it as it's moving through the barrel. Some sort of circular disk to minimize escaped air (say, made out of an old mouse pad?) first placed at the bottom of the barrel with the ball then inserted so that it's lying on top of it might improve performance. But in the end I've decided that all I really need to do is add a few more psi with my bike pump to get the heights I need.

Resources:

Here (An excellent site!)
2" ID launcher

(Googling "spud gun," "potato gun, and "tennis ball launcher" will provide other sites with great ideas, too.)

Caveats:

If you build one of these, use common sense and, above all, use at your own risk! Follow instructions for gluing PVC, allow adequate curing time, and, when finished, don't overstress the PVC by pumping in too much air!

Solid-stating the Heathkit HR-10 Receiver

2011-05-04T09:30:00.000-07:00

In a previous post I detailed my experiences in modifying a Heathkit HR-10B receiver. Although performance improved, I've never been entirely satisfied with those mods, mainly because, to my ears, there is a very subtle distortion that seems to occur with loud signals. I suspect that the input into the NE602 product detector stage is a bit too high (because the AVC isn't doing a great job limiting signal levels?), and the oscillator is being pulled slightly on high-level signal voice-peaks.

Rather than continue to incrementally modify that HR-10B receiver to improve its performance, I thought an interesting project would be to completely solid-state an HR-10 ( or HR-10B). However, I didn't want to rip the guts out of the HR-10B that I was currently using -- it was in too nice a condition, physically, for that (which is why my mods that I'd made to it can be easily backed-out).

Luckily, I found a junker HR-10 receiver that was exactly what I was looking for: rusty, almost complete, inexpensive, and looking for a home -- the perfect playground for experimentation!

Here's the top of the chassis, as received. Just a wee bit of oxidation...

(Click on image to enlarge)

And here's the bottom of the chassis. Everything looks like its there!

(Click on image to enlarge)

I started by removing all of the parts except those I expected to use (e.g. the RF transformers, Oscillator tank components, and the variable caps), sanded the chassis to remove the oxidation, and then I began designing, building, and testing...

The final receiver chassis: rust removed (via sanding), modifications installed, and ready to receive signals!

(Click on image to enlarge)

Schematics:

Here are the schematics for the new receiver:

Page 1: RF Input

This page contains the Input RF Filters, the first Mixer, and its VFO. It uses the original RF bandpass components (L1-L5 and their associated capacitors) as well as the Oscillator tank components (L11-L15 and their associated capacitors).

(Click on image to enlarge)

Notes on Page 1:

All components with reference designator values less than 100 are original HR-10 components.
Replaced the antenna connector with a BNC.
I couldn't get good performance using the NE602's internal oscillator with the existing HR-10 oscillator tank circuits, so I designed a separate oscillator using a J310.
The 8.2 ohm resistor and ferrite bead the Q101's gate ostensibly prevent VHF oscillations, but I've not verified if they really do any good, or not.

Page 2: IF Filter, IF Amplifier, and AVC

This page contains the IF Filter, the IF Amplifier, and the AVC circuitry.

(Click on image to enlarge)

Notes on Page 2:

All components with reference designator values less than 100 are original HR-10 components.
There is roughly 20 dB of loss through T1 and the crystal filter, which Q201 compensates for (plus a few dB).
The original loads for the MC1350 had been just the 330 uH inductors, but the circuitry was unstable. Adding 5.1K resistors in parallel with each inductor calmed it down. I didn't bother to try it with just the 5.1K resistors as loads.
D201 prevents the AGC (aka AVC) voltage that drives the MC1350's AVC control pin from exceeding the MC1350's power supply.
One output of the MC1350 drives the SSB demodulator (single-ended). The other output is used to derive AGC from the 1.68 MHz IF signal. Thus AGC is IF-derived, not audio-derived.
But first the IF signal is amplified by Q202 and Q203 before it is rectified by D202. (This amplified signal will also be used as the source for AM demodulation on page 3).
Similar AGC voltage results were achieved whether D202 was a silicon, Schottky, or germanium diode. So I left the diode as a silicon one.
C217 provides an RF "ground" for the AGC reference rail (U202.8), and R224 helps isolate the op-amp's output from any high-frequency IF signal (or rectified IF signal) that might appear on this rail (via the AVC cap, for example).
The input of the MC1350, when driven single-ended, cannot exceed 2.5 Vpp or else distortion occurs at its output.
As the output of the MC1350 driving the NE602 SSB demodulator (on page 4) is increased from about 20 mVpp, the audio signal becomes more and more distorted (although this distortion might be difficult to hear). For example, given a signal generator's signal that's been tuned in by the receiver so that it produces a 1 KHz audio signal at the speaker, if the level of the IF signal from the MC1350 is 30 mVpp, the audio second harmonic (2 KHz) is about 40 dB down from the fundamental. If the MC1350 output is increased to about 100 mVpp (by reducing AGC loop gain), the second harmonic increases to be only 20 dB down, and there is noticeable "pulling" of the BFO oscillator frequency. For this reason I set the AGC loop gain (via R228) to keep the MC1350 output level at about 30 mVpp so that the second harmonic was 40 dB down from the fundamental. There is some IF signal overshoot (to about 50 mVpp) when a -30 dBm signal goes suddenly from OFF to ON, but there is no noticeable audible "popping" at the speaker from the overshoot. (Note: overshoot worsens as R228 (AGC Loop Gain) is decreased in value -- this also corresponds to an increase in output level from the MC1350 and increased harmonic distortion at the demodulated audio output, as previously discussed.)
With D202 a silicon diode and the gains set by the component values shown in the schematic, AGC action doesn't start to limit a signal (on 80 meters) until the input signal level reaches about -100 to - 90 dBm. From that point, the AGC Voltage (at pin 5 of the MC1350) increases in 0.03 volts steps (roughly) for each 10 dB step in input signal level until the input signal reaches about -30 dBm, at which point the input stage (NE602) limits the signal. AGC Voltage varies from about 3.92 volts (no signal) to 4.12 volts (input limiting).

Page 3: Demodulation and AF Amplification

This page contains the SSB and AM demodulators and the AF Amplification chain.

(Click on image to enlarge)

Notes on Page 3:

All components with reference designator values less than 100 are original HR-10 components.
Q301, when ON, connects the BFO tank to ground so that the BFO can oscillate. R301 provides a DC path for the Q301's collector (as there is no DC path through T5) to ensure that the transistor is always ON.
In SSB mode, C306 provides a pole at about 5 KHz (with the NE602's output resistance of 1.5 Kohms). And for both SSB and AM, C308 provides an additional pole at about 8 KHz.
Q302 provides additional amplification of the IF signal so that it can drive the AM detector consisting of diodes D301 and D302 (the lower this signal is, the more clipping occurs on the "low" side of the modulation envelope because the signal doesn't exceed the diode turn-on thresholds).
C318 compensates for the crystal filter's passband shape (which results in a low-frequency "hump" in the audio when operating AM). Adding a zero at about 1 KHz (C318 = 1N, R318 = 150K) reduces this hump, thus flattening the AM passband so that it sounds less bassy.
C320 adds a pole at about 4 KHz when in AM mode, helping to reduce the "hiss" of the wideband noise from the AM Detector (detecting noise from the MC1350 output).
The LM1875 came out of my junkbox. Other audio amps should work fine, too.

Page 4: S-Meter and Calibrator
This page contains the S-Meter and 100 KHz calibrator circuitry.

(Click on image to enlarge)

Notes on page 4:

All components with reference designator values less than 100 are original HR-10 components.
The S-Meter amplifier has a gain of about 16, which was a compromise. Because of the AGC control-voltage characteristics which are used to drive this meter (see discussion for page 2 of the schematics), if the gain was set so that the needle was at S9 for a -73 dBm signal, then, as the signal level was increased by 10 dB, the needle would move by 20 dB on the S-meter scale and it would quickly peg on the right side. Conversely, if the gain was set so that the needle for an S-9 + 60 dB signal was at the far right meter tick and the needle moved by 20 dB for a 20 dB change in signal level, the needle sat at about S3 when there was no signal. The problem is that the AGC voltage operates over a smaller range of signal levels than those represented by the meter scale (in which S0 is -127 dBm, S9 is -73 dBm, and S9+60 dB is -13 dBm). So I threw up my hands and compromised with the values shown.
The 7490 Decade Divider in the calibrator circuit is NOT wired to produce a 100 KHz square wave. Rather, it's wired to generate a 100 KHz signal whose duty-cycle is 20% so that even harmonics, as well as odd harmonics, are produced (a square-wave with a duty-cycle of 50% produces no even harmonics!).

Page 5: Power Supply

This page contains the power-supply and dial-light circuitry.

(Click on image to enlarge)

Notes on page 5:

All components with reference designator values less than 100 are original HR-10 components.
The power supplies should be self-explanatory. To remove 120 Hz hum (and its harmonics) from the 17V rail (for low-noise applications), I used a simple filter consisting of R501 and C506.
The 1815 bulbs are rated at 200 mA each for 14 Volts. Lifetime is 3K hours.
I use a string of 8 diodes (rather than a resistor) to drop 17 VDC down to something lower for the lamps -- diodes will keep the lamp voltage constant even when bulbs with different current draws (e.g. 1813 or 756) are used in lieu of the 1815 bulbs. Fewer diodes can be used, but the 8 diodes in series give me a brightness I was satisfied with. The voltage across the bulbs is dropped to about 11 volts, and this lower voltage should increase bulb life. At 11 volts the two lamps, together, draw about 0.32 Amps total (measured through R510), which means that each diode dissipates about a quarter-watt each (or 2 watts, total).

Construction:

After removing most of the original HR-10 components, I started building up my new circuitry on sheets of PCB material that I'd screwed to the chassis:

(Click on image to enlarge)

I used my own construction technique, which is simply mounting components so that I can read their values. ICs are mounted facing up, and I usually fold out their pins (so that they look like wings) and mount them by soldering a couple of the pins of each IC to components mounted vertically on the PCB (e.g. a bypass cap (on the power pin) or to a 1 Megohm resistor that is standing up with one end tacked to the PCB copper plane (you need to first ensure, though, that 1 Meg to ground will not affect the signals using that pin!)).

Other 1 Meg resistors (I have a large reel of them here) are soldered vertically (one end to the copper sheet) to serve as mounting posts for other components. In my opinion, this method is easy and it beats trying to solder or glue little pads made of PCB material to the copper sheet (one technique used by others).

A closeup of my construction technique:

(Click on image to enlarge)

Yes, I know. It isn't pretty. But it works.

Additional Notes and thoughts:

1. SSB versus AM passbands

With the SSB passband adjusted (via T1) to be fairly flat, the passband in AM mode was very narrow -- noticeably less than 2 KHz, and thus AM signals sounds pretty bassy.

In order to get a bit more frequency range in AM mode, I readjusted T1 to make the AM frequency response fairly flat out to about 2 KHz. However, this put a 7 to 8 dB hump (in LSB mode) at the high end of the audio spectrum.

The plots below show this. The grey graph is the frequency response to noise in AM mode, while the blue graph is the frequency response to noise in LSB mode. (Noise fed to the antenna connector from an external RF noise generator).

(Click on image to enlarge)

Yes, the hump looks terrible, but during listening tests I didn't find it to be too objectionable on LSB, and so, for the moment, I've decided to keep these passbands as they are (as a compromise between AM and SSB), but I might change my mind in the future. (Note, too, that this hump appears as a bass hump in USB mode).

2. MDS Levels, by band:

By ear (rather than quantitatively), MDS (Minimum Discernible Signal) on the different bands is roughly the following:

80 meters: -130 dBm
40 meters: -130 dBm
20 meters: -120 dBm
15 meters: -90 dBm
10 meters: -110 dBm

As you can see, both 15 and 10 meters are pretty deaf. I've not yet found a solution for this problem, and, because I don't spend any time on these bands, this is not a very high priority for me. However, there does seems to be a bit of VFO blow-by on these two bands which is getting into the AGC detector (and thus adding attenuation to the signal path), which isn't helping. I added a shield between the VFO coil assembly and the MC1350 (consisting of a piece of copper-clad PCB material mounted vertically and soldered to ground) which seems to help reduce this VFO-pickup, but it hasn't cured the problem when 15 meters is selected.

Also, both 15 and 10 meters hetrodyne the signal using the second harmonic of the VFO. If the VFO is clean (i.e. it looks like a sine wave) there will be very little harmonic content and this could affect the conversion gain. Unfortunately, the conversion gain of the NE602 is directly related to the VFO signal level (up to a point), and therefore, in the case of 15 and 10 meters, if the amplitude of the VFO's second harmonic is low, so will be the resultant IF signal, which is why it can sound deaf on those bands (I verified this, by the way, using an external generator as a VFO. With its frequency set to the original VFO's second harmonic (e.g. 22.78 MHz to receive 21.1 MHz)-- sensitivity on 10 and 15 meters improved at VFO amplitude levels comparable to those used for 80 and 40 meters.)

One possible solution might be to add a frequency doubler to the output of the VFO for 15 and 10 meters to increase the amplitude of the second harmonic. We'll see...

3. VFO Frequencies, per band:

80 meters: 5.18 - 5.68 MHz (VFO = F + 1.68 MHz)
40 meters: 8.68 - 8.98 MHz (VFO = F+ 1.68 MHz)
20 meters: 15.68 - 16.08 MHz (VFO = F+ 1.68 MHz)
15 meters: 11.34 - 11.565 MHz (VFO = (F+ 1.68 MHz) / 2)
10 meters: 14.84 - 15.69 MHz (VFO = (F+ 1.68 MHz) / 2)

4. Image Rejection:

On 80 meters I can sometimes hear 40-meter shortwave broadcast stations. For example, if the VFO is tuned to 5.53 MHz (to receive a 3.85 MHz signal), the receiver will also pick up a signal at about 7.21 MHz, which is the image of the 3.85 MHz signal (5.53 MHz + 1.58 MHz). A -70 dBm signal at 7.21 MHz is only about 20 dB down from a -70 dBm signal at 3.85 MHz -- not very good image rejection. An external antenna tuner (low pass topology) can improve this rejection, though.

One way to improve image rejection might be to add an RF preamp prior to the NE602 and use the existing resonant L/C circuits from the original HR-10 RF Preamp (L6-L10). However, some amount of attenuation would probably need to be added, too, so that NE602 isn't overdriven by loud signals. Currently, on 80 meters, it starts to conk out at around -30 dBm, and for that reason I wouldn't want to add additional gain prior to the input NE602, unless this gain is counterbalanced with an equivalent loss.

5. Oscillator Drifts:

There is some amount of drift when the receiver is first turned on, but it seems to stabilize fairly quickly.

On 80 meters, overall receiver drift, from a power-off state, was measured to be roughly 1000 Hz in the first minute. Three minutes later it had drifted another 400 Hz, and from then on it settled down to an overall drift on the order of +/- 50 Hz over an hour.

To separate out VFO drift from BFO drift, when the VFO was replaced by a Fluke 6060A signal generator, the BFO drift, from a power-off state, was measured to be about 300 Hz over one hour.

6. ANL Switch: I haven't yet wired up the ANL switch (nor designed ANL circuitry) -- it's a feature I rarely use, and in the future I might decide to assign a different function to this switch (e.g. selectable input attenuation, or...?).

7. Other oddities:

On 10 meters you can pick up the 17th harmonic (loud!) of the BFO (at around 28.56 MHz).

Future Improvements:

Someday...

1. Improve performance on 10 and 15 meters (possibly by adding a frequency-doubler circuit to the VFO for these two bands?).

2. Add an ANL circuitry (for the existing ANL switch).

3. More RF filtering to improve image rejection.

4. Add REC/STBY function to octal connector on rear of chassis so that can mute receiver if used with a transmitter.

Caveats:

1. I could have easily have made a mistake, so please regard (and use) this design accordingly.

2. I make no claims that component values are the optimum ones which could be used -- rather, I used values and components which, from the data-sheets and my design equations, seemed to be appropriate choices, and I modified these as needed. The values I've used work for me, but I've not spent any time evaluating the design from the perspective of "optimal" (rather than "good enough") component selection.

Class E/F Exciter for the 813 AM Transmitter

2011-03-29T14:30:00.001-07:00

This exciter replaces the Johnson Ranger that I'd originally used to drive my 813 AM Transmitter (described here, here, and here). It uses a modern Class E/F PA (described in further detail here), and it has a separate audio amplifier to drive the modulator deck in the 813 rig.

(Click on image to enlarge)

There is one major difference, however, between my original Class E/F PA, which was designed to generate 40 watts of RF power, and this final PA. This difference commences with a big...

Oops!

For when I connected this exciter to the 813 rig and keyed it for its initial "smoke test," the 813 Transmitter's grid-current meter pegged.

Oops!

But this shouldn't be! I'd measured the power output of the Ranger when it was driving the 813 Transmitter, and this output was around 40 watts for me to drive the 813 rig to about 350 watts. My exciter put out the same power. What was going on?

As an experiment, I took a 50 ohm 6 dB high-power attenuator (that had been wired-in under my operating position) and connected it between the new exciter and the 813 PA's RF input. When I keyed the rig, the PA's grid current rose to about 22 mA -- right around where it was when the Ranger was driving the rig.

Hmmm...

I poked around and discovered that the 6 dB attenuator I'd just tested with had originally been installed between the Ranger and the 813 transmitter. I'd forgotten about it, and I'd assumed that the Ranger had been directly driving the 813 rig with 40 watts, when in reality it had been driving the 813 transmitter rig with one-quarter of this power! Doh. Dope slap!

An obvious solution was to keep the 6 dB high-power pad connected between my exciter and the 813 PA Deck's input, but this seemed like a waste of a good attenuator (high-power attenuators are expensive, after all). Was there another, simpler, way to decrease the output power of my exciter by a factor of 4?

A Slight Change to the Design...

The Exciter's voltage for the FET Drains was 26 volts. If I halved this value, then, in principle, I ought to get a quarter of the power (power changes with the square of voltage).

Luckily, the design already has a 12V switching regulator (rated to 3 Amps), so I just moved the connection for the FET Drain power from 26V to the output of the 12V switching regulator. Keyed it up, and, voila, it worked! The meter readings for the 813 were where they were when driven with the Ranger.

Length Matters...

One interesting phenomena that I noticed when doing this, though: during my initial testing, I'd connected the Exciter's RF output to the PA's RF input through two lengths of RG-58 coax (because I'd originally placed the 6 dB pad between these two lengths) for a combined length of about 9 feet. Later, when I shortened the total coax between the Exciter and the PA Deck from about 9 feet to 3 feet, the PA Deck's Grid Current started reading in the 35 mA range rather than in the original 20 mA range and the Exciter's Drain current jumped from about 0.8 A to 1.2A. Neither of these were desired changes, so I went back to my original 9 feet of coax to interconnect the Exciter to the PA Deck.

Why does a change of 6 feet make such a difference in operation? At the moment, I don't know. However, as the length of the interconnecting coax is shortened, Exciter Drain current increases (from about 0.8A with 9 feet of coax to about 1.2A with 3 feet of coax), so the implication is that, with shorter coax, the Exciter is seeing a lower load resistance. This then implies that the PA Deck's RF input doesn't look like 50 ohms resistive, and thus there is an impedance transformation taking place via the 50 ohm coax.

(I connected an HP 3577A network analyzer to the exciter-end of the coax feeding the PA Deck. With the PA grid tuning set to peak grid current (when the exciter was connected), I made the following measurements:

9' coax: S11 mag: 0.92, S11 angle: -13.2
4.5' coax: S11 mag: 0.67, S11 angle: -24.1

When converted into a parallel representation of real and imaginary impedance components (because, after all, the Exciter's tank consists of parallel-connected components, not series), the resulting values are:

9' coax: Real: 47.4 Ω, Imaginary: -j202 Ω
4.5'coax: Real: 36.8 Ω, Imaginary: -j82 Ω

Assuming that the Exciter tank is tuned to compensate for the imaginary component, the Exciter tank sees a lower resistive component with shorter coax, which correlates with the increased Drain current that I see, and the resistive component with 9' of coax is quite close to 50 ohms.

However, there is one puzzle that I don't yet understand: with 4.5' of coax, the imaginary component represents more parallel capacitance than that of the 9' coax, yet I find that, when tuning the Exciter's tank when using the 4.5' length of coax, I need to turn the Exciter's Tank capacitor to full-mesh (i.e. high-capacitance) for best-looking Exciter RF. Why do I need to add more exciter-tank capacitance when I've already added more capacitance at the exciter load? It doesn't make sense to me. Should I be working with the series-form of impedance instead (in which the impedance measured at the end of the 4.5' length of coax has less capacitance than the 9' length)? Have I made a mistake in my measurements? I don't know.

Well, something to research on another day...

Other notes:

Note 1: If I'd kept the 6 dB attenuator connected between the Exciter and the PA Deck, then the effect of coax-length on Exciter performance would be less of an issue, because the attenuator would have "buffered" the effect of the PA Deck's input impedance on the Exciter.

Note 2: There is interaction between the Exciter's Tuning capacitor and the PA Deck's Grid Tuning capacitor; the position of one will affect the other. That is, the amount of "junk" on the Exciter's RF waveform (monitored at the front-panel BNC) will change, depending upon how the Grid Tuning capacitor is changed. When tuning the transmitter:

First I peak the PA Deck's Grid current.
Then I adjust the Exciter Tank tuning for best looking RF at the Exciter's output (as observed at the Exciter's front-panel BNC). This is typically at, or near, minimum Drain current, as measured on the Exciter's front-panel meter.

Here's a screen-shot of bad-looking RF from the Exciter. Its tank needs tuning!

(Click on image to enlarge)

And here's the Exciter RF with its tank properly adjusted:

(Click on image to enlarge)

(Exciter RF waveform measured at the front-panel BNC, J6, using a Tektronix TDS320 scope (100 MHz bandwidth).)

Schematics.

There are four pages. Here they are:

(Page 1. Click on image to enlarge)

Notes on page 1:

This page is essentially the same as the original design, but changes are:

DC Voltage for IRF530s changed from 26 VDC to 12 VDC.
510 pf added to tank circuit (3570 pf total) so that the Tank circuit, when operating at 3.87 MHz, is properly tuned with Tuning Capacitor C11 at about half-mesh.

(Page 2. Click on image to enlarge)

Notes on page 2:

No change from the original circuit. But because the LM2576 switching-regulator now must deliver an additional 800 mA (or so, to power the PA FETs), the inductor L3 really should be changed from 1000 uH to 470 uH or 330 uH. But it seems to run fine with the original value of 1000 uH, so I'll leave modifying this for another day.

And, strictly speaking, I didn't need to incorporate a sequencer into the Exciter's design -- I could have used the existing sequencer in the 813 transmitter to perform the same function. But incorporating this sequencer allows me to easily test the Exciter as a stand-alone unit.

(Page 3. Click on image to enlarge)

Notes on page 3:

This is the audio driver which drives the 813 Modulator Deck. Externally, and prior to this stage, I use a Behringer Xenyx 802 mixer/amplifier to amplify and equalize my microphone.

For 100 percent modulation, the Modulator Deck requires an input level of about 80 volts RMS (when driven with a sine-wave -- this is about 226 Volts peak-to-peak). The simplest way to get this sort of amplitude is with a transformer. On eBay I found an audio output transformer (designed to present to a push-pull driving stage a load of 6.6K or 8K ohms when driving either a 4, 8, or 16 ohm load -- its Part Number is OT20PP), and I decided to connect it in reverse to drive my Modulator Deck so that I could transform the high-impedance of the Modulator Deck input to a low-impedance, and then drive this low-impedance with a speaker amplifier designed to drive loads in the 4 to 16 ohm range.

To test which combination of input/output windings would work best in my application, I connected the transformer to the Modulator Deck and drove it with a stereo amplifier. With a 1 KHz sine-wave test signal, for full modulation (corresponding to an audio drive of about 80 Vrms into the Modulator deck), I needed about 12 Vpp of drive from the stereo amp.

For the actual transformer driver, I used an LM1875 speaker amplifier. Its output is single-ended, so, to get 12 Vpp out with some headroom, I used the 26 VDC power supply to power it.

I also decided to use the 16 ohm tap as the primary (driven by the LM1875) and the 6.6K ohm taps as the secondary (to drive the Modulator Deck). This is the lowest step-up turns-ratio provided by the windings, and the Modulator Deck's input impedance is transformed to be about 5.4 ohms, as measured at the output of the LM1875, which conveniently lies between the LM1875's 4 ohm and 8 ohm load specs. (Any other combination of windings would have resulted in a lower load impedance for the LM1875).

When driving the Modulator Deck to full modulation, the LM1875 delivers about 3.6 watts into this 5.4 ohm load.

There is a potentiometer to allow some amount of gain adjustment, but the primary gain is back at the Behringer mixer. And there's a mute circuit to mute the audio drive to the Modulator Deck when the 813 Transmitter is not transmitting. (The 813 transmitter does not like it when the modulator and modulation transformer are driven when the PA deck is not generating RF).

The low-frequency -3 dB point is about 280 Hz (determined by R28 and C44), which I purposefully added when I discovered that lowering this frequency caused the AM signal to sound a bit fuzzy (due to IMD products related to the voice frequencies below this point). The upper -3 dB point for the exciter/813 rig (combined) is around 4 KHz. These points were measured by driving the modulator with sine-waves and measuring the peak-to-peak envelope of the modulated RF.

As a precaution against EMI problems involving RF interacting with the audio components, the audio components are all placed within a separate shielded chamber (made using double-sided PCB stock) within the chassis. All signals which transition into this chamber from the area containing the Exciter's RF stages are first filtered using feed-thru caps and L/C (or R/C) low-pass filters.

(Page 4. Click on image to enlarge)

Notes on page 4:

The 26 VDC power supply is a Cosel 24V supply (adjustable +/- 10%), rated at 4.5 ADC that I picked up from eBay. Now that I've discovered that I don't need 40 watts of RF power, this supply could actually be rated at a much lower DC output current, but hey, hindsight is 20/20.
The AC Connector and AC Line filter are actually an integrated modular unit.
The VFO is an N3ZI DDS2 VFO. Its output is only about 380 mVpp, so I bump it up to about 2 Vpp (to drive the 'HC86 XOR gates) with an OPA690 op-amp. The 50 ohm resistor in series with the output was added to reduce some high-frequency ringing I had observed, but I'm not sure it's really needed -- I may have been mistaken in this measurement.
The VFO Amp is only turned-on when transmitting. With the chassis buttoned-up, I've found that, even though the DDS VFO is always active (even during receive), I cannot hear it on my receiver.
The Drain Current meter is 1.5 mA full-scale. The resistors (and sense-resistor) scale the current reading so that the meter represents actual current ÷ 2000.
For adjusting the Tank's tuning capacitor, I added an RF tap (R32 and R33) which connects to a BNC on the front panel. The series-2K ohms represented by R32 and R33 help to isolate the tank circuit from the capacitance of coax-cables used to connect this BNC to a scope.
And a diplexer is still used to help clean up the Exciter's RF output. The 50 ohm load for the Diplexer's parallel L-C circuit (i.e. the load for out-of-band frequencies) is actually seven 357 Ω, 1/4 watt resistors in parallel. And I placed the series L-C part of the diplexer in a Pomona box because I was concerned that, if not shielded, unwanted RF components would couple around it to the output.

813 Transmitter Wiring Diagram with the K6JCA Exciter installed.

(Click on image to enlarge)

And here are some photos!

The Audio stage. Note the shielded compartment. And the LM1875 amplifier attached to the side of the chassis for heatsinking.

(Click on image to enlarge)

In the rack and on the air!

(Click on image to enlarge)

(Note: This shot was taken with the FETs powered with 26V, rather than 12V, and a 6 dB attenuator between the Exciter and the PA Deck. With a 12V FET power source, the meter needle is about 0.4 mA (out of 1.5 mA FS), representing about 0.8 Amps of Drain current.

Additional Notes:

Because the tank transformer is 1:1, I wondered what the effect would be if I moved the Tuning Capacitor (C11) from the primary side of the tank to the secondary side. This would allow me to more easily mount the cap, because it no longer would need to float. However, when I performed this experiment, I discovered two issues:

The tuning range narrowed.
Output power varied slightly with frequency.

Neither of these outcomes were positive, so I kept the tuning cap on the input side of the tank transformer, and I mounted it on a piece of polycarbonate plastic (from Tap Plastics) to isolate it from the chassis.

Resources:

Datasheets:

Caveats:

1. I could have easily have made a mistake, so please regard (and use) this design accordingly.

2. High voltages can kill. Use caution.

TX Overshoot on Flex 5K

2011-03-03T08:46:00.001-08:00

[Additional Notes:

Latest News, 9 MARCH 2011:

Per FLEX, this problem is now FIXED!!!

Gerald's Email to FlexRadio Customers states, in part:

"Fixed ALC overshoot and corrected leveler gain
target in the transmitter audio signal chain.
These changes have been verified by customers
on air and in the FlexRadio lab using a digital
storage oscilloscope to eliminate overshoot."

7 MARCH 2011:

The word from Flex is that they've tested a solution which looks very promising.

Also, additional analysis reveals that the overshoot isn't due to Gibbs Phenomena (as I first hypothesized (see below)), but instead I now believe it's because ALC action occurs before the "real" TX signal is converted to I and Q. Shifting a signal's frequency components each by 90 degrees can result in a time-domain signal whose amplitude exceeds that of the original ALC-limited signal, as demonstrated in the graph below:

(Click on image to enlarge)

Blue Waveform (Sawtooth)
= sin(2πx) + 0.5(sin(2π2x)) + 0.33(sin(2π3x)) + 0.25(sin(2π4x)) + 0.2(sin(2π5x)) + 0.167(sin(2π6x))

Red Waveform (Sawtooth w/components shifted 90^o)
= cos(2πx) + 0.5(cos(2π2x)) + 0.33(cos(2π3x)) + 0.25(cos(2π4x)) + 0.2(cos(2π5x)) + 0.167(cos(2π6x))

[By the way, despite the way it looks, the Red waveform does not have a DC shift. If you examine the equation from which this graph was generated, you'll note that there are only cosine terms in the equation, and that there is no "constant" term, which would represent a DC shift.]

Imagine that the Blue waveform, being our TX signal, had been "normalized" by ALC operation to have a maximum amplitude of 1.0 (i.e. divide all the values in the graph above by 1.5 so that the Blue waveform has a Vpeak of 1.0). The Red waveform, which is the TX signal with its frequency components shifted by 90 degrees, would have a peak amplitude exceeding 1.0 if its components were also normalized by the same amount (divided by 1.5).

And so, to properly limit the signal, ALC must be applied after the conversion from real to complex data. Which, coincidentally, my fix (described below) does.

Interestingly, if a Triangle-wave's components are phase-shifted by 90 degrees, the resultant signal has an amplitude less than the original's amplitude. Which correlates nicely with our observation that we don't see overshoot with a Triangle wave as our signal, but we do if the signal is a Sawtooth wave.

My original post is below...]

On the topic of TX overshoot with the Flex 5000...

Last summer a friend of mine mentioned that he was having a problem with PowerSDR and his Alpha amplifier -- it looked to him as though RF overshoots at the Alpha's input were causing it to shut down, and these shutdowns were occurring frequently enough to really annoy him.

I decided to do a bit of investigation. When I looked at the RF envelope from my Flex 5000, I saw overshoots with my 1.18 Console. Thinking that the problem might be related to the software revision, I downloaded the 2.0.7 PowerSDR Console and ran my tests again. Below are snapshots of the OUTPUT RF waveform from my 5000 using the 2.0.7 console.

Oscilloscope is my Tek 2445 that I use to monitor my transmit RF (via an RF-sampler). The two horizontal cursors you see on its CRT mark the peak-to-peak level of the RF signal when the 5000 is in TUN mode and the Tune level is set to 10 watts.

For the following snapshots, PowerSDR is set to:

Freq: 3.863 MHz
Mode: LSB
Drive: 10
Leveler: Disabled
TX EQ: OFF
DX: OFF
CPDR: OFF
DEXP: OFF

(Why use 10 watts and not 100? At some point the PA will limit and it won't be able to deliver any additional power. If we run the experiments at high power, we risk having PA-limiting influence our results. But if we run our experiments at low power, we should be able to get a better representation of how high the peaks actually go, because they won't be subjected to PA limiting.)

Anyone can try this experiment with these setting. Here are my results:

First, two nicely behaving waveforms:

Notice how the peaks don't exceed (by much) 10 watts?

But look at these next two. Yikes!

The four snapshots are:

Triangle waveform from internal internal software Transmit generator. Looks great!
Pulse, from the internal software Transmit generator (at its default settings for Pulse mode). Looks great!
Sawtooth waveform from the internal software Transmit generator. Yikes -- it greatly exceeds the 10-watt cursors! Peak power is 49 watts on my LP-100 power meter, yet DRIVE is set to 10 watts.
My voice, saying "Ahhh" loudly into microphone. Yikes again! Peak power is about 40-50 watts on my LP-100 power meter, yet DRIVE is set to 10 watts.

Both the "Ahhh" and Sawtooth signals greatly exceed the 10 watt Drive setting on the 2.0.7 console, and the level of the voice signal is very similar to what I'm seeing with my modified 1.18.4 console.

In AM mode, the sawtooth (as modulation) looks exactly as it should, so I don't believe the issue with the sawtooth is that it's somehow "broken."

I thought I'd experiment with the code a bit and see if I could gain further insight...

First, I experimented with the ALC code itself, as my first suspicion was that the overshoot was due to the non-zero attack time. But nope, that wasn't it. I changed the attack time to 0 (so that the ALC was a true peak-detector, and not one with a non-zero attack time), and it made no appreciable difference in the gross overshoots experienced by voice or sawtooth SSB modulation.

I then played around with the location of the ALC algorithm, and the results are very interesting. If the ALC code is prior to the filter_OvSv routine (as it is in the version of code I'm using (v1.18.4) as well as in the SVN 3862 code that I downloaded), then both Voice and Sawtooth waveforms grossly overshoot the target power, per the photos above. In other words, the current ALC code in its current position, in either my console (1.18.4) or in the 2.0.7 console, does not properly limit the output RF for certain waveforms.

However, if I move the ALC code to just after the filter_OvSv routine (and change the buffers used for ALC in newDttSPAGC from buf.i to buf.o), then there is minimal overshoot for all waveforms (Tone, Triangle, Sawtooth, and Voice), and their peaks are all near the target power. In other words, the output RF looks beautiful. [Note: I wasn't able to test the PULSE waveform, because my 1.18 version of the console doesn't have PULSE as one of the options for the test generator].

I'll make a wild guess that the overshoot is due to the bandpass filtering in filter_OvSv and thus similar to Gibb's phenomena. But this is just a guess. Conceptually, it makes sense to me that the ALC processing should be as late in the processing chain as possible, and moving it to just after filter_OvSv seems to be a better place for it than prior filter_OvSv. However, I'll be the first to admit that I don't know what the ramifications are of doing this. Are other problems introduced? I don't know.

Building an 80-Meter Class E/F RF Amplifier...

2011-01-29T13:05:00.000-08:00

After simulating on my computer an 80-Meter Class E/F amplifier (here and here), I decided to actually build one. Here it is:

(Click on image to enlarge)

With a 26V power supply, RF Power Out is about 40 watts. And efficiency of the MOSFET final is in the range of 85-95% (depending upon which power meter I use to measure RF power).

In the image above:

The two MOSFETs (IRF530, N-Channel) are mounted on the vertical copper plate at the upper-left.
The Tank circuit, including the transformer (5-turn air-wound coil) and a variable air capacitor (for tuning the tank) are at the upper-middle.
Control logic (and Symmetry adjustment potentiometers) are at the middle-left.
Two solenoid-style inductors (MOSFET Drain DC feeds) are lower middle.
A 12V switching voltage-regulator is at the lower-right (with the large toroidal inductor).
You can also see the two scope-probes attached to the MOSFET Drain busses, and the RF output is the BNC at the right.

Here are the schematics:

(Click on image to enlarge)

Notes on the Design:

Page 1:

This page contains the MOSFET Amplifier and its driving circuitry.

The amplifier consists of a pair of IRF530 MOSFETs in a push-pull configuration that drive a tank circuit consisting of transformer T1 and parallel capacitors C7 and C10. Why IRF530 MOSFETs? They were in my junk box!

These MOSFETs are rated at 100 volts max V_DSS and have an R_DS(on) of about 0.18 Ω (the latter depends upon which manufacturer's datasheet you look at).

The tank transformer, T1, consists of two windings, 5 turns each, with the secondary winding wound inside the primary winding (the windings are concentric). Total coil length is about 2.75", and the inner-diameter of the outer coil is about 1.5". After I wound T1 I discovered that its inductance measured to be around 570 nH. I'd been shooting for about 400 nH, but the difference isn't a big deal -- 570 nH just lowers the overall Q a bit (from a simulated Q (with 400 nH) of 5 to an actual Q of 3.6 at 3.87 MHz with a 50 ohm load). (See note later in this posting regarding measuring inductance of an unknown inductor).

C7, which, in combination with C10, forms the resonant tank capacitance, is actually six 510 pf ceramic capacitors (low ESR caps from American Technical Ceramics (their 700B series)). C10 is an air variable (20-420 pf) from my junk box, and it gives me a tuning range of approximately 3.78 MHz to 4.03 MHz.

Peak voltage across these tank components is on the order of 80 volts or so, so there's no reason for high-voltage parts. However, current through the inductor and capacitors is on the order of 3 to 5 amps RMS (per my SPICE simulations), so low-ESR components are highly recommended.

The MOSFET Drains are fed via 18 uH inductors L1 and L2. I had wanted to use higher inductance, but I didn't have anything in the junk box that was suitable (high inductance and high self-resonant frequency (S.R.F.)). However, I did have a couple of J.W. Miller 5252 inductors -- these are 125 uH inductors, but their self-resonant frequency is only about 2.5 MHz. I removed the top two winding layers (of three total layers) to give me an inductance of 18 uH and an S.R.F. of about 48 MHz.

I decided that the simplest way to drive the PA MOSFETs would be with MOSFET drivers. I chose IXDD414 MOSFET drivers (note: these are obsolete parts, but I found mine on Ebay). They drive the MOSFETs via 1-ohm resistors, which seemed to reduce ringing (but this observation really should be reconfirmed -- take it with a grain of salt).

To drive the IXDD414 Drivers I use two XOR gates to generate, from the VFO signal, two signals of the same frequency but 180 degrees out of phase with each other. One of the XOR gates inverts the VFO signal, while the other passes it through uninverted. Both XOR gates are on the same die, which should minimize the differential delay between the two signals.

The VFO signal is AC-coupled to an input on each of these two XOR gates, and two potentiometers provide variable DC offsets to each of these same two inputs so that the duty-cycle (i.e. symmetry) of each XOR output can be independently adjusted.

(Note: For testing I drove the XOR gates with an HP 8640B signal generator, set to +10 dBm. At this level, this generator provides a nice, very low distortion sine-wave with about a 4 Vpp amplitude (rather than 2 Vpp, which you'd expect at +10 dBm, because the 8640B is now terminated in a high impedance, rather than 50 ohms)).

The final bit of circuit on this page of the schematic, Q1 (a P-channel MOSFET), ramps the DC voltage feeding the IRF520 Drains up and down when the amplifier is turned on and off, thus providing a soft, rather than hard, transition to the output RF envelope. (I tried using the enable pins on the IXDD414 ICs in lieu of adding this MOSFET, but I found there were voltage spikes on the Drains of the IRF530 MOSFETs exceeding their V_DSS when I transitioned these Drivers ON using their enable pins).

Page 2:

This page contains the voltage regulators and control circuitry.

An LM2576 switching regulator provides the +12VDC power for the IXD414 MOSFET Drivers. These two drivers require about 0.4 - 0.5 amps of current, total. A 12V linear regulator would have had to dissipate about 5 to 6 watts of power, which is why I chose to go with a more efficient switching regulator.

The values for the switching-regulator's components are straight out of the LM2576 datasheet. Note that this datasheet specifies a 1000uH inductor for the 12V regulator when the input voltage is around 26V and the load current around 0.4 Amps. Pulse Electronics has a series of "50 KHz Inductors" (available through Digikey) that are recommended for LM2576 applications. I used the PE-53120, which is a 1000 uH inductor.

A 7805 5-volt linear regulator provides 5VDC for the digital logic.

This amplifier is designed to be keyed by my 813 AM transmitter. Because I want the VFO to be inaudible in my receiver when I'm not transmitting, I need some way to disable it (or move it off frequency) when I'm not transmitting. Also, I want the VFO to be enabled and generating its signal before I apply power to the MOSFET Drains and to go OFF after I remove power from the MOSFET Drains when I'm done transmitting, so that the VFO is stable at all times while power is being applied to the MOSFETs. In other words, I want to "nest" the MOSFETs' ON/OFF cycle within the VFO's enable/disable cycle.

Although I could use the 813 Transmitter's sequencer to nest the MOSFET Power ON/OFF within the VFO Enable/Disable, for ease-of-testing I decided to incorporate a sequencer into this design to allow me to test this amplifier as a stand-alone unit. This new sequencer is also based upon the W2DRZ design. In the future, when I incorporate this circuit into the 813 AM Transmitter, I will decide if I should use the 813 AM Transmitter's sequencer in lieu of this one.

This sequencer runs at a 50 Hz clock rate because I want it to run faster than the sequencer in my 813 Transmitter (which I've set to run at roughly a 10 Hz clock rate).

I plan to drive this amplifier with an N3ZI DDS Module, which I would like to disable when I'm not transmitting so that it doesn't interfere with reception. There are a couple of ways that this might be done. One is to shift the VFO to a completely different frequency (using the DDS Module's "VFO B/A" input).

Moving the VFO off frequency when not transmitting would prevent Receive interference, but it might be also useful to shut off the VFO when transmitting so that the MOSFET Drivers (which are driven by the VFO via the XOR gates) aren't consuming power from the 12V supply. There are several ways in which one might do this:

One way might be to "lift" (via an open-collector/drain driver) the ground-end of R1, the 6.8K resistor attached to the AD9834's FS ADJUST pin (pin 1), when not transmitting. Per the AD9834 datasheet:

IOUT_{FULL SCALE} = 18 * FSADJUST/R_SET
(FSADJUST = 1.15V nominal, and
R_SET is R1 in the N3ZI DDS2 VFO schematic)

Thus, raising the resistance of R1 during Receive should reduce IOUT to near zero.

Another possible way to disable the VFO might be to simply short the IOUT output from the AD9834's IOUT to ground. This is a current-source output, of which the DDS chip has two (IOUTB is the second one), and the datasheet states that IOUTB can be shorted to ground if not in use, so I would think that one could also short-out the first output, too, to disable the DDS (both are rated at the same output current). But is this advisable? I can't say. Also, the current at IOUT would still be generated and thus sourced out the IOUT pin, and so any "non-zero area" current-loop formed by the "shorting" components and IOUT's signal path could still create some amount of interfering RF emissions on the receive frequency.

And perhaps the easiest way would be to simply take the IXDD414s' EN pins (pin 5) to ground at the same time that the VFO frequency is shifted (i.e. VFO disabled).

There are some signals on this schematic page, such as MUTE VFO, that could be used for just this purpose (e.g. tie MUTE VFO to the U8.5 / U9.5 node). I'll determine which route is best when I incorporate the N3ZI DDS module.

Notes on Construction:

I built the circuit on a piece of 6.5" x 4.5" scrap single-sided copper-clad FR4 circuit board. The copper plane on this board is used as the circuit ground. I cut up pieces of double-sided FR4 PCB material to use as mounting pads and power busses (one side of each is soldered to the copper "ground plane" on the main board).

Regarding the MOSFETs and their MOSFET drivers (and any other components with high slew-rate signals), it's important to minimize parasitic inductance, so keep leads as short as possible.

Caps used for power-supply bypassing (e.g. C23-C26, C29 and C30) should have very high self-resonant frequencies, and they should be mounted as close to the IXDD414s power pins (and ground) as possible to minimize unwanted inductance from their leads.

It's a good idea to try to ensure that each Drain of the IRF530 MOSFETs sees approximately the same amount of capacitance-to-ground from the wiring and other sources -- my SPICE simulations showed that unequal amounts of capacitance-to-ground for the two MOSFETs results in unequal voltage peaks, with respect to ground, at the Drains of the MOSFETs.

The IRF530 MOSFETs are attached to a heatsink consisting of a copper buss-bar that's 1" x 4.75" x 0.125". This heatsink is electrically attached to the circuit ground with copper tape and also a small angle bracket that mechanically holds the heatsink to the circuit board.

The transformer T1 is constructed of rectangular 3/16" x 1/16" enameled copper magnet wire (this is the same wire used by Taniguchi, Potter, Rutledge in their 200 W Power Amplifier which appeared in the Jan/Feb 2004 issue of QEX). It consists of two windings, 5 turns each. Total length is about 2.75", and the inner-diameter of the outer coil is about 1.5". The secondary-winding is closely wound inside the primary winding (to minimize leakage-inductance), and I covered one of the windings with Kapton tape when I found that the wire's enamel was sometimes cracking as I bent it, and I was concerned that the two windings might short-out to each other.

The LM2576 datasheet has useful tips for layout and interconnection of components.

Adjustments and Tuning...

There are three adjustments to make: The two Symmetry potentiometers, R3 and R6, set the duty-cycle of the outputs of the two XOR gates, and the Tune variable capacitor, C10, in the output Tank Circuit allows the Tank's resonant frequency to be tuned roughly 250 KHz.

Initial Setup:

While not transmitting (PTT IN is still high), I first set the two pots, R3 and R6, to midway in their adjustment range. While monitoring the XOR outputs with a scope, I then I tweak the two potentiometers so that the outputs of the XOR gates are each at (or close to) a 50% duty-cycle.

I then take PTT IN low to enable the transmitter. While monitoring the two Drain waveforms on an oscilloscope, I tweak the two pots to try to minimize the "ringing" on the MOSFET Drain signals. (See note below regarding how to accurately probe the Drains). Note: The final setting of each pot seems to correspond to a duty-cycle very close to 50% for each XOR output.

Then I adjust the frequency of the VFO for minimum Drain current (using a voltmeter across the 0.1 Ω resistor, R12, to monitor DC Drain current).

I again tweak the pot adjustments to again minimize ringing on the two Drain waveforms. (If you have a Spectrum Analyzer, you can also monitor the spectrum (out to, say, 100 MHz) and adjust the pots to minimize the higher-frequency spikes).

The result should look something like this:

(Click on image to enlarge)

Once these adjustments have been made, I find that I can keep the pot settings fixed and only change C10 when I move to a different frequency. (When moving to a different frequency, you can try retuning C10 for minimum Drain current, but I personally find that approach only gets me into the ballpark. The approach I follow is described in more detail just a bit later in this section.

Per my measurements, the efficiency of the MOSFET amplifier itself is around 90% (this number should be taken with a grain of salt, as it depends a great deal upon the accuracy of your watt-meter as well as DC current and voltage measurements!), and efficiency drops to about 86% at about 100 KHz of either side of the frequency to which C10 is tuned to. However, ringing is starting to look pretty severe at this point, and so I'd recommend changing the VFO by no more than, say, +/- 50 KHz before retuning C10.

When misadjusted, the Drain waveforms can look like the following image (or much worse!). Note that this is ringing on the waveform, not oscillation.

(Click on image to enlarge)

When changing frequency, I only adjust the capacitor, C10. The two pots I leave alone after their initial adjustment (see above).

When tuning to a new frequency, my procedure to adjust C10 is a bit iterative. You can try to adjust it for minimum Drain current, but I find that this approach doesn't always work. Instead, I do the following:

Rotate C10 to where you think it should be, approximately, for the frequency you'll be using.

Tune the VFO for minimum Drain current.

If the frequency of the VFO, after tuning, isn't close enough to where you'd like to be, tweak C10 a bit and repeat.

This approach seems to work for me.

Knocking Down the High-Frequency Junk

The output of the Class E/F amplifier is fairly dirty with high-frequency trash from 1) Harmonics (and ringing) on the MOSFET Drain waveforms, and 2) pickup of harmonics from the XOR gates and MOSFET Drivers.

Here's the spectrum of the Amplifier's output prior to adding an external filter.:

(Click on image to enlarge)

(Measurements made through a 30 dB Bird attenuator. The signal at the far left is the fundamental (at about 3.8 MHz), and the next signal (very low level) is the second harmonic.)

The above chart shows spurs out to 100 MHz. In reality they extend well past this frequency. In an attempt to reduce them, I first tried a 5-element Chebychev low-pass filter from the tables in the ARRL Handbook (Fig. 12-19, #7, for f_co of 4.5 MHz, which results in values of 620 pf for the input and output caps, 1200 pf for the middle cap, and 2.39 uH for the two inductors).

But when I tried using this filter, I had to crank up my power-supply to about 30 volts to get the same amount of RF power output. So I nixed this filter (The peak Drain-Source voltage for the IRF530 MOSFETs was at its limit).

Instead, I decided to incorporate a simple diplexer, similar (in topology) to the one described in the Part 2 article of the High-Efficiency Class-E Power Amplifiers. . To keep the design simple, I designed for a Q of 1, which means that the inductors and the capacitors are the same in both "halves" of the diplexer. Here's its schematic:

(Click on image to enlarge)

The caps are dipped silver-mica (300V rating), and the inductors were wound with 26 gauge enameled wire. I'm using a BNC-mounted Tek 50 ohm, 1/2 watt termination to terminate the parallel L-C branch of the diplexer. After about 1/2 hour of "key-down" operation, this termination gets a little bit warm.

With this filter, the PA requires a DC Power Source of about 25.7 VDC to generate 39 watts RF Power out.

And here's the resulting spectrum:

(Click on image to enlarge)

Because of its low Q, the Diplexer doesn't do much to knock down harmonics that are near the fundamental, such as the 2nd and 3rd harmonics. But because of the symmetrical output of a push/pull amplifier topology, the 2nd harmonic is already quite far down (when the Symmetry pots are properly adjusted), and the 3rd harmonic is now more than 40 dB below the fundamental.

One advantage to a diplexer implementation is that it presents a near 50 ohm load to all frequencies (assuming that the external load presented to the series L-C filter is 50 ohms at its resonant frequency). This can help stabilize an amplifier that might otherwise be unstable when presented with off-frequency unknown impedances.

Accurately Probing the MOSFET Drain Voltage Waveforms

If you try measuring Drain voltage waveforms with a standard scope probe and its ground lead, you are not going to get an accurate picture of what the waveform actually looks like. Instead, you are very likely to see all kinds of junk on the signal. This "junk" really isn't on the signal -- it's radiated noise picked up by the inductive "loop" formed by the scope probe and its ground lead.

To get an accurate picture of how the signal really looks, you need to minimize the capture area of this loop. The best way to do this is to keep the ground "lead" as short as possible, and connect ground to the probe as close to the probe tip (where the signal being probed is) as possible.

For this purpose I adapted a Tektronix PCB to probe-tip adapter (Tek p/n 131-4244-00, made for probes such as the P6139A). I soldered two pins of the four-pin "ground" shell to the PCB copper-clad upon which I'd built the amplifier (the other two pins are in the air, as you can see in the photo below), and the socket for the probe tip I soldered to the Drain buss for one of the MOSFETs. The other MOSFET has the same setup so that I can monitor both Drains simultaneously.

(Click on image to enlarge)

(There's a useful app note from Analog Devices here. Although it discusses probing high-speed signals, its techniques are quite applicable to this circuit, too.)

Other Notes and Thoughts:

1. Measuring Inductance or Self-Resonant Frequency of an inductor:

Here's the technique I use. There might be better approaches, but this one worked for me...

(Click on image to enlarge)
2. Tank Capacitors:

Per my SPICE simulations, the tank circuit has large currents -- on the order of 3 to 5 Amps RMS. So you want the tank capacitance to have a very low ESR to minimize unwanted power dissipation.

For my tank capacitance, I used six 510 pf caps in parallel. The capacitors are manufactured by American Technical Ceramics, and are from their 700B series. The ESR of their 510 pf capacitor is in the range of 0.04 - 0.05 Ω. If we assume 0.5A of current passes through each cap (ball-parking 3A RMS total current through six caps), the power dissipation in each is then on the order of 0.01 watts. Not too bad!

(In hindsight, though, I probably should have used their 470 pf caps. The 510 pf caps have a working voltage of 100 volts, which is a bit too close to the voltage they're actually seeing in this circuit (see scope images above of the Drain waveforms). The 470 pf caps, on the other hand, have a working voltage of 200 V, which gives a nice margin. If I can get these caps in, I'll replace the 510 pf caps.)

3. IRF530 MOSFETs: These have a max V_DSS of 100V. I'd like to have a bit more margin (per the discussion above regarding the 510 pF capacitors), but that's what was in the junk box when I looked for suitable parts. We'll see if there's any issue with reliability.

4. Some useful formulas:

For a resonant circuit: L = 1 / [ (2*π*f)² * C ], or C = 1 / [ (2*π*f)² * L ]
For a series RLC circuit, Q = (1/R) * √L/C, which, when combined with either of the two equations above, gives us: Q = |X/R|.
For a parallel RLC circuit, Q = (R) * √C/L, which, when combined with either of the two equations above, gives us: Q = |R/X|.
Note that when Q = 1, R = |X| for both the series and parallel RLC circuits.

5. 24V vs. 12V Power Supplies:

This design could be run from 12V (e.g. 13.4 VDC) rather than 24V (actually 26V) by changing the design of T1 to transform the 50 ohm load to a lower resistance across MOSFET Drains. This approach has two advantages: 1) Lowers the peak-voltage across the Tank components and at the Drains of the MOSFETs, and 2) Eliminates the need for the 12V switching regulator.

However, with the 24V supply, I can adjust the supply's output voltage (typically +/- 10%) to vary RF Output Power independently of the 12V supply (in case there are other radios running on the same 12V supply that might depend upon that voltage being fixed). Also, use of a 24V supply means that the transformer T1 for this application has a turns-ratio of 1:1, which (in my mind) simplifies its design. It would need to be on the order of 1:4 for a 12V application.

6. Transformer T1:

T1 consists of two concentric solenoid coils, air-wound with thick wire to minimize losses. It's possible that this transformer could be wound on toroidal or balun forms instead, using smaller gauge wire than the very large rectangular magnet wire that I used. It would be interesting to see what the effect of such a transformer would be on overall efficiency. Perhaps someday...

7. RF Power and Efficiency versus Power Supply DCV:

(Click on image to enlarge)

Notes:

DC Voltage measured at the Q1 side of L1 and L2.
The burbles in the efficiency curve are probably due to measurement errors on my part.

References:

Datasheets:

Misc:

W2DRZ Sequencer
N3ZI DDS2 Module
Analog Devices App Note: High-Speed Time-Domain Measurements -- Practical Tips for Improvement
Toroid Winding Site

Spice:

Free Download of LTSpice here.
LTSpice "Getting Started Guide" here.

Class E Amplifiers:

Sokal, "Class-E RF Power Amplifier", Jan/Feb 2001, QEX
Sokal, "Class-E High-Efficiency RF/Microwave Amplifiers", (a more detailed paper)
Lau, Chiu, Qin, Davis, Potter, Rutledge, "High-Efficiency Class-E Power Amplifiers" Part 1, May '97, QST, and Part 2, June '97 QST
Davis, Rutledge,"A Low-Cost Class-E Amplifier with Sine-Wave Drive"
Der-Stepanians, Rutledge, "10-MHz Class-E Power Amplifiers"
Melia, Robert, O'Conner, Class-E Power Amplifier Design
Tayloe, Class E Amplifiers (Norcal QRP Presentation)
Class E Design Software from Tonne Software (I've never used this, so use at your own risk)
WA0ITP Class-E Amplifier Design Spreadsheet (I've never used this, so use at your own risk)
Class-E AM Forum
Class-E AM Transmitters (WA1QIX)
Class-E Amplifier Experiments , Calculations, and Notes on Designing Class-E RF Amplifiers, all by Bill Slade. (I've not verified the accuracy of these posts.)

Class E/F Amplifiers:

Taniguchi, Potter, Rutledge, "A 200 W Power Amplifier", Jan/Feb 2004, QEX
Letters, May/June 2004, QEX
Letters, July/August 2004, QEX
Kee, Aoki, Hajimiri, Rutledge, "The Class E/F Family of ZVS Amplifiers"
Kee, "The Class E/F family of Harmonic-Tuned Switching Power Amplifiers" (Cal Tech Thesis)
Kee, Aoki, Rutledge, "7.1 MHz, 1.1 KW Demonstration of the New E/F2,odd Switching Amplifier Class"
Jeon, "Design and Stability Analysis Techniques for Switching Mode Non-Linear Circuits: Power Amplifiers and Oscillators" (Cal Tech Thesis)
Kee, et al. U.S. Patent, No. 6,724,255
Niknejad, "Class E/F Amplifiers" (presentation, EECS 242, U.C. Berkeley)
Bohn, Kee, Hajimiri "Demonstration of a Switchless Class E/Fodd Dual-Band Power Amplifier (a 40 and 30 meter dual-band amp)

Caveats:

1. I could have easily have made a mistake, so please regard (and use) this design accordingly.

Modeling Class E/F RF Amplifiers, Part 2

2011-01-27T05:17:00.000-08:00

[Part 1 can be found here]

In Part 1 I evaluated a Class E/F amplifier using a center-tapped transformer to supply power to the drains of the two active devices.

DC power can also be fed to the drains of the two MOSFETs via two inductors, thus simplifying the transformer because it no longer needs to be center-tapped. I was curious how such a circuit compared (in performance) to the circuit described in Part 1.

Here's the SPICE model (using Linear Technology's free SPICE program: LTSpice):

(Click on image to enlarge)

Regarding the model:

1. C4, at 1 Farad, provides a "stiff" AC ground for simulation purposes so that there's no voltage fluctuation at the R4, C4, L1, L2 node.

2. L1 and L2 provide a high-impedance feed for DC power to the "Drains" of the MOSFETs. I chose these to be (initially) 25 uH.

3. The MOSFETs are modeled with voltage-controlled switches. Their Ron is set via external resistors to be 0.15 Ω (plus 0.01 Ω within the Switch model).

4. 500 pf capacitors mimic the MOSFETs' C_oss (per the IRF 530 datasheet). Note that these capacitors are not voltage dependent.

5. Refer to Part 1 for more detail on selecting the Transformer inductance of 400 nH.

6. C3 was arrived at by trial-and-error (after initially setting its value to 3.655 nF per Part 1) by adjusting its value so that the voltage waveform at node "Va" is zero when the current waveform (through R1) is non-zero.

(Click on image to enlarge)

When adjust the value of C3 (or the transformer's inductance), the current and voltage waveforms maintain the same relationship to C3's value (or to the transformer's inductance) as was described in Part 1. Refer to the image below.

(Click on image to enlarge)

Lowering RMS current through the switches:

I thought I'd see if I could lower power dissipation in the switching devices by lowering their RMS current (see Part 1 for more discussion on this technique). By lowering the value of inductance of L1 and L2, I could replicate current waveforms through the switching devices that look as though the amplifier is now a a "Class E/F_2,odd" amplifier, but, when I made my measurements using the tools in LTSpice, there was really only minimal effect (if any) in power dissipation.

Here's how the waveforms look with L1 and L2 reduced to 800 nH. As the table following this image shows, although the current-waveform changes significantly, there isn't much of a difference in switching-device power-dissipation (as measured across R1 and R2).

(Click on image to enlarge)

I used the LTSpice measurement functions to measure power and RMS current at different points in the circuit (for various values of L1, L2), and these measurements have been tabulated below.

(Click on image to enlarge)

Some observations from this table:

I really cannot see much change in R1's power dissipation as L1, L2 are varied. That is, the differences are small, and perhaps are within the tolerance of simulation errors.
Also, there's not much change in either Output Power (Pload) or efficiency for the range of values of L1 and L2 shown in the table. Therefore, a L1, L2 values in the range of, say, 25 uH - 100 uH, might be the simplest approach for my application.
As L1, L2 are decreased, C3 must be increased to compensate for their effect on the time-relationship of the voltage and current waveforms. However, an increase in the value of C3 increases the current through C3 (because voltage across the transformer is essentially constant, and we decrease C3's reactive impedance when we increase its value). Therefore, due to its internal ESR, C3 will dissipate more power when L1,L2 are reduced.
As the values of L1 and L2 is increased, it takes longer for the simulation waveforms to ramp up to their final values.

From the minimal change in R1 dissipation, and from the gross changes in C3 current (and thus, potentially, its power dissipation), it really doesn't make much sense (in my application) to attempt to "tune" L1 and L2 to minimize power dissipation in the switching devices by minimizing their RMS currents.

It's possible, though, that in applications in which higher currents are present, some benefit might be achieved by "tuning" L1 and L2. However, the simulations don't bear this out in my application, and there seems to be little benefit.

Overall, the power-out from this circuit topology is comparable to that of the center-tapped transformer topology described in Part 1. So, for my application, either circuit should do the job, and it'll just be a matter of determining which one is easiest to build.

Modeling with the IRF530 MOSFET

Actual MOSFETs can be simulated via LTSpice's "nmos" model, within which one can retrieve the SPICE parameters for a number of different MOSFETs, including the IRF530. (Note: these devices are not alphabetized in the "Pick New MOSFET" table.)

Below is a circuit with IRF530 MOSFETs substituting for the original voltage-controlled switches. The final value of C3, as well as the AC source's amplitude and offset voltage (V2), were arrived at by trial-and-error. Also, two AC sources are used (the second with an offset of 180 degrees), because it seemed like an easy way to provide two sine-wave sources that were 180 degrees out of phase but with the same DC offset.

(Click on image to enlarge)

There are several differences between this circuit model and the original one with the voltage-controlled switches:

The Sine Wave sources have an amplitude of 10 Vpeak and a DC offset of 5V (to bias the IRF530's near their turn-on point). This gives peak gate voltages of +15/-5 Volts (note that the max rating of the IRF530's gate voltage (V_GS) is +/- 20V).
C3 is now 3.69 nF.

Here is the Drain Voltage versus Source Current for one of the IRF530 MOSFETs:

(Click on image to enlarge)

Note: Source current is shown, rather than Drain current, because the Drain pin has quite a bit of current flowing through it even when the MOSFET is OFF. This current at the Drain pin is not dissipative current, though (that is, it isn't being dissipated as I² * R heat through R_DS(on). Instead, it's current passing through the Gate-Drain capacitance and out the Gate pin (LTSpice plots verify that I(drain) - I(gate) = I(source)).

Here are some of the currents and powers (and overall efficiency) measured via LTSpice.

RF Power Out: 48.2 watts
DC Power In: 52.0 watts
Efficiency: η = 48.2/52.0 = 93%
I_L1 = 1.14 A_RMS
I_C3 = 4.5 A_RMS
I_L3 = 5.2 A_RMS

Note the high currents in C3 and L3. Care should be taken to use low-ESR devices!

References:

Spice:

Free Download of LTSpice here.
LTSpice "Getting Started Guide" here.

Class E Amplifiers:

Sokal, "Class-E RF Power Amplifier", Jan/Feb 2001, QEX
Sokal, "Class-E High-Efficiency RF/Microwave Amplifiers", (a more detailed paper)
Lau, Chiu, Qin, Davis, Potter, Rutledge, "High-Efficiency Class-E Power Amplifiers" Part 1, May '97, QST, and Part 2, June '97 QST
Davis, Rutledge,"A Low-Cost Class-E Amplifier with Sine-Wave Drive"
Der-Stepanians, Rutledge, "10-MHz Class-E PowerAmplifiers"
Melia, Robert, O'Conner, Class-E Power Amplifier Design
Tayloe, Class E Amplifiers (Norcal QRP Presentation)
Class E Design Software from Tonne Software (I've never used this, so use at your own risk)
WA0ITP Class-E Amplifier Design Spreadsheet (I've never used this, so use at your own risk)
Class-E AM Forum
Class-E AM Transmitters (WA1QIX)
Class-E Amplifier Experiments , Calculations, and Notes on Designing Class-E RF Amplifiers, all by Bill Slade. (I've not verified the accuracy of these posts.)

Class E/F Amplifiers:

Taniguchi, Potter, Rutledge, "A 200 W Power Amplifier", Jan/Feb 2004, QEX
Letters, May/June 2004, QEX
Letters, July/August 2004, QEX
Kee, Aoki, Hajimiri, Rutledge, "The Class E/F Family of ZVS Amplifiers"
Kee, "The Class E/F family of Harmonic-Tuned Switching Power Amplifiers" (Cal Tech Thesis)
Kee, Aoki, Rutledge, "7.1 MHz, 1.1 KW Demonstration of the New E/F2,odd Switching Amplifier Class"
Jeon, "Design and Stability Analysis Techniques for Switching Mode Non-Linear Circuits: Power Amplifiers and Oscillators" (Cal Tech Thesis)
Kee, et al. U.S. Patent, No. 6,724,255
Niknejad, "Class E/F Amplifiers" (presentation, EECS 242, U.C. Berkeley)
Bohn, Kee, Hajimiri "Demonstration of a Switchless Class E/Fodd Dual-Band Power Amplifier (a 40 and 30 meter dual-band amp)

Caveats:

1. These simulations are for educational purposes only.

2. I could have easily have made a mistake, so please view (and use) these simulations accordingly.

3. And a final caveat regarding SPICE modeling. Modeling results should always be taken with a bit of caution, for the ability of SPICE to mimic actual circuit behavior depends in large part on how accurately a circuit and its components have been modeled. Factors such as parasitic components and circuit non-linearities can all cause modeled SPICE performance to diverge from actual performance. SPICE can provide valuable insights into circuit operation, but a bit of skepticism, too, should be applied when evaluating results.

Modeling Class E/F RF Amplifiers, Part 1

2011-01-26T13:32:00.000-08:00

[Part 2 can be found here.]

I've recently been thinking of replacing the Johnson Ranger which drives my 813 AM transmitter with something more modern and which consumes less power -- a homebrew, solid-state unit which would replace the Ranger's RF driver and Speech Amplifier.

The Ranger drives my 813 AM transmitter with 36-37 watts of RF Power(SWR is 1.27:1, by the way). I have a DDS VFO module that I'd like to use as the frequency-control of my Ranger-replacement, but its output is probably on the order of milliwatts (not yet measured), so I'll need to boost it with (probably) several stages of RF amplification.

With that thought in mind, I started researching possible amplifier circuits that might provide the required 40 - 50 watts of RF power on 75 meters.

I'd like to maximize efficiency of the PA, and of course Class-C Amplifier topologies immediately leapt to mind. But while researching designs, I came across the Class-E mode of amplification which promised improved efficiencies over Class C. This looked very worthwhile, and the design procedures for such amplifiers seem to be well established.

However, as I searched the literature, I stumbled across an interesting QEX article ("A 200 W Power Amplifier", in the Jan/Feb 2004 issue) describing a Class E/F amplifier for 40 meters. The Class E/F topology (being push/pull) offers benefits over Class E, such as:

Lowers the required Vdc power supply voltage for the same power out.
Lowers Vds across the MOSFETs for the same power out.
Improves harmonic suppression.

Further investigation was needed!

Unfortunately, design methodology for the Class E/F amplifiers doesn't seem to be at the same state as that of Class E amplifiers. Not letting that deter me, I figured SPICE modeling of these amplifiers would allow me to play around with circuit parameters and component values, and I would (hopefully) gain some insight into their operation and possible component values for my application. (By the way, there's a free SPICE program available via the Linear Technology website).

Building a SPICE Model of a Class E/F amplifier:

Below is my SPICE model for a Class E/F_odd amplifier. It's topology is based upon the amplifier circuit described in the 7 MHz, 1.1 KW demonstration amplifier by Kee, Aoki, and Rutledge). My goal was to design a PA for 75 meters (3.87 MHz, typical frequency) and output power (at the 50 Ω load resistor) on the order of 40-60 watts.

(Click on image to enlarge)

(Note: Comments are in BLUE, SPICE directives are in BLACK).

How did I arrive at the component values in the above circuit? Let's look at my procedure:

First, my givens: Rload = 50 Ω. Desired power out: 50 watts.

Next, I wanted to select a power-supply voltage (Vdc) that would:

Allow me to use a 1:1 impedance transformation from load to the differential connection across the PA MOSFET drains.
Not be too high of a voltage so that I could use 100V (V_DSS) MOSFETs that are in my junk-box (e.g. IRF 530) .
Yet be high enough to minimize current through the MOSFETs and the resultant "i² R" power loss through their R_DS(on) resistance.

Per my design goal of an amplifier that delivers 50 watts into a 50 Ω load, this means that the voltage across the 50 Ω load is 70.7 volts peak (141.4 Vpp or 50 volts RMS).

And per my goal of not requiring impedance transformation (either from high-to-low or low-to-high impedance) from the load to the MOSFETs' drains, I'm using a 1:1 transformer to present the 50 Ω load resistor as 50 Ω to the PA MOSFETs' drains (this load appears differentially between the two drains).

From the Class E/F literature, the peak voltage across the drain of each MOSFET is ≈ π*Vdc, where Vdc is the power-supply voltage (refer to equations 37 & 38 in "The Class E/F Family of ZVS Switching Amplifiers", for example). So, in my case of an untransformed (i.e. 1:1) 50 Ω load, then the peak value of the voltage at the drain of either MOSFET is the same as the peak value across the 50 Ω load resistor. In other words, Vds(peak) = Vload(peak), where the load is R3 in the circuit model above. Using this equality, we can derive the following equation:

Vdc ≈ [ √(Pload * Rload) ] / [ π * (√2) / 2]

Where Pload = power delivered to Rload (R3), and therefore Pload = [ (√2 / 2)*π*Vdc]² / Rload, using our definition of Vpeak = π*Vdc.

In this case in which Power = 50 watts and Rload = 50 Ω, Vdc calculates to be 22.5 volts. (Note: I'm using 22.8 volts for my modeling, which is the lower-limit of the output voltage adjustment of some off-the-shelf 24 volt switching supplies), and Vds (peak) across either MOSFET is π*Vdc, or 71.6 volts peak. The IRF 530 MOSFET has a rated V_DSS of 100 volts, so there's almost 30 volts of margin.

OK, now it's time to calculate the inductance that we'll need between the drains of the two MOSFETs. This is the inductance of the primary of the center-tapped 1:1 transformer. Because the transformer is 1:1, the inductance of the primary is the same as the inductance of the secondary (same number of turns in each). To determine the inductance of the primary, let's first assume it has a loaded Q of 5. (I'm actually going to use 5.1). Why 5? In fact, some of the Class E/F amplifiers in the literature use a Q of around 3 or even less. I'd like to keep Q sort of high so that harmonic suppression isn't too compromised, but this may need adjusting if/when I actually build the circuit.

OK -- the inductance of the transformer's primary is loaded by Rload (R3), reflected through the transformer from the secondary. Thus, because the transformer is 1:1, this 50 Ω load appears as 50 Ω across the primary, and it is this 50 Ω that "loads" the Q.

Because L and R are in parallel, we know that:

Q = |R / X_L|

In this case Q is actually Q_loaded, and so we can rewrite this equation to give us L:

L = R/(2 * π * f * Q_loaded),

which, in our example, gives us the following:

L = 50/(2*3.14*3.87e6*5.1)
= 400 nH

Now let's use this value to model our transformer...

To create a center-tapped transformer in LTSpice, we need to use three coupled-inductors. Two of these inductors represent the turns on either side of the primary center-tap (and thus each has half the number of turns of the secondary, given that the transformer is 1:1), and the third inductor is the secondary.

We know that the primary should have an inductance of 400 nH. Therefore, the secondary, because it has the same number of turns as the primary, will also have an inductance of 400 nH.

The two inductors representing the two halves of the primary each has half the number of turns as the secondary. Therefore each has 1/4 of the inductance of the secondary (inductance has a turns-squared relationship). So we specify each of these two inductors, in SPICE, to have 100 nH inductance. This gives us a 1:1 center-tapped transformer. (Refer to LTSpice "Help" for more information on Transformer modeling, and an example of a model for a 1:3 transformer).

Let's give these inductors some series resistance so that they represent "real" rather than "ideal" components -- that is, they have some amount (albeit small) of series resistance. The two inductors in the primary, because they're coupled, represent 400 nH overall inductance. Let's assume an unloaded Q of 150 for this overall inductance. Because this is a series R-L configuration, we use a slightly different formula for Q:

Q = |X_L / R|

Solving for R:

R = 2 * π * f * L / Q
= 2*3.14*3.87e6*400e-9/150
= 0.065 Ω

I've rounded this down to 0.06 Ω, and I'm splitting it into two 0.03 Ω resistors which I place in series with either side of the transformer's primary.

(Note: in the past (and perhaps still) SPICE programs sometimes had problems converging upon a solution if one specified only ideal components. From habit I tend to throw in small values of resistance in series with an ideal component, such as the 0.06 Ω in series with the inductor(s) described above, with the hope that they'll nip such problems in the bud.)

What should be the value of the resonating capacitor (C3)? Unfortunately, I couldn't find an equation for calculating this value. All I know is that the overall impedance of the parallel L-C circuit represented by the transformer primary and C3 (in the circuit above) should be slightly inductive (per Class E/F references). This means that, because the circuit is a parallel-resonant circuit, the resonant frequency of the L-C combo should be slightly above the circuit's operating frequency.

But how far above should it be? I don't know, and I cannot find an equation to calculate it. [For further discussion of how one might calculate C3, please refer to Note 3 in the Notes section at the end of this posting].

So perhaps we should run some simulations and see if we can find it by trial and error...

But before I get to that, let me first fill in the remaining components in my SPICE circuit model:

I've set the dc-feed series inductor (L1) to an arbitrarily large value, in this case 10 uH (for no good reason, I chose 25 times 400 nH).

C4 is set to the humongous value of 1 Farad to provide a "stiff" AC ground. (At 1 uF, the value I'd originally set it to) there was some voltage fluctuation at the C4, L1, R4 node, and so I thought it better, for simulation purposes, to make the AC ground here as solid as possible).

I'm modeling the MOSFETs with (almost) ideal voltage-controlled switches (LTSpice's "SW" model). Because I'm not concerned (at the moment) with modeling the MOSFET inputs and drive circuit, I can define the inputs to be whatever is most convenient for me. Thus, I've set the switching thresholds of the SW model to 0 volts, and the switches are driven by a 1-volt amplitude sine-wave that has a 0 volt DC offset. Each switch is on for one-half of a full cycle, and neither switch is on at the same time.

The default SW model in LTSpice has an Ron of 1 Ω, which is much too large for our purposes. The IRF 530 MOSFET that I'd like to use has an R_DS(on) of 0.15 Ω (depending upon which manufacturer's datasheet you look at). Although I could specify Ron in the SW model to be 0.15 Ω, I personally prefer to specify it as a separate component in series with each switch, and thus I also changed the SW model's Ron value from 1 Ω to an arbitrarily small value (in this case 0.01 Ω).

Similarly, I've placed the MOSFETs' Drain-Source Output Capacitance (C_oss) across each series combination of 0.15 Ω and SW model. In the case of the IRF 530, C_oss is 500 pf, which I've shown here in my circuit model.

So now we're back to C3. How do I determine its value? Let's use SPICE to find it by trial-and-error...

Let's start by assuming it resonates with 400 nH (the transformer's primary inductance) at our operating frequency of 3.87 MHz. This gives us a value for C3 of 4.22 nF.

Plugging this value into the model and running a simulation, it's quite apparent from the large current spikes that the voltage waveform at the "drain" of S1 is misaligned with respect to the current flowing through S1):

(Click on image to enlarge)

Playing around with the value of C3 and iterating to a solution, I found that a value of 3.655 nF for C3 gives a good result for the voltage and current waveforms. (This also results in a resonant frequency of 4.16 MHz).

(Click on image to enlarge)

Experiments!

Now that we've created our SPICE model of a Class E/F amplifier, let's do some experiments.

First, let's try decreasing and increasing C3 (and thus raising and lowering, respectively, the resonant frequency from the original value of 4.16 MHz).

Lowering the resonant frequency to 4.08 MHz by increasing C3 to 3.8 nF gives us this relationship between the voltage and current waveforms:

(Click on image to enlarge)

So if we see waveforms that looks like the above, we know that we need to decrease the capacitance of C3 to shift the voltage waveform to the left, relative to the current waveform.

Now let's raise the resonant frequency to 4.25 MHz by decreasing C3 to 3.5 nF:

(Click on image to enlarge)

And if we see waveforms that look like the above, we know we need to increase the capacitance of C3 to shift the voltage waveform to the right, relative to the current waveform.

Notice in the above images how the voltage waveform shifts with respect to the current waveform as we move the L-C network's resonant frequency up and down. Our goal is to adjust C3 so that we minimize the power dissipated within the MOSFETs (Vds is 0 volts when current is non-zero). This leads us to the following rule-of-thumb, as expressed in the image below:

(Click on image to enlarge)

Lowering RMS current through the switches:

Reduction of the RMS current through the switching devices can be accomplished by resonating L1 with the switch's C_oss (C1 or C2) near the second harmonic of the operating frequency (refer to section "C" in the 7 MHz, 1.1KW Amp reference for more information). This type of configuration is known as a "Class E/F_2,odd" amplifier.

Here's the resulting waveforms when L1 is changed from 10 uH to 500 nH.

(Note: C3 had to be changed from 3.655 nF to 3.798 nF to bring the voltage and current waveforms back into their proper time-relationship.)

(Click on image to enlarge)

A couple of points:

1. I played around with different values of L1 (modifying C3 each time to maintain the proper time relationship for the Voltage and Current waveforms) and measured, via LTSpice measurement functions, RMS current and power at different points in the circuit. Here's a table of my results:

(Click on image to enlarge)

Some observations from this table:

Current (RMS) through a switch (i.e. I(R1)) is minimized when L1 = 500 nH. BUT, the delta in power dissipation (from, say, the case in which L1 = 100 uH) is only about 0.02 watts.
Also, current through other devices increases as the value of L1 is decreased. There is no "sweet spot" for these other currents when L1 = 500 nH, and the other devices will dissipate more power (because of their higher currents) as the value of L1 is decreased to 500 nH.
Overall, there's not much change in either Output Power (Pload) or efficiency for the range of values of L1 shown. Therefore, using an L1 value in the range of, say, 10 uH - 100 uH might be the best approach for my application.
I can't explain why there's as much difference as there is when L1 = 240 nH.
As C3 is increased to compensate for a decrease in L1, C3's current increases. Therefore, due to its ESR, it will dissipate more power. (However, the delta in current when going from an inductance of 100uH to 500nH isn't too bad.)
As the value of L1 is increased, it takes longer for the simulation waveforms to ramp up to their final values.

2. L1, at 500 nH, resonates with the 500 pF value of C1 at about 10 MHz, which, although near to the second harmonic of 3.87 MHz, is actually a bit closer to the third harmonic. I don't know if there's a formula one can use to exactly calculate what L1 should be -- the literature seems to only state that the resonant frequency should be "near" the second harmonic. Iteration through trial-and error might be the only approach (at this time) for selecting an inductance for L1 to minimize switching-device power dissipation.

3. From these results, it's doubtful that, for my application, tuning L1 for minimum I_RMS, through the switches would be worth the effort. However, this approach might prove more worthwhile in applications in which significantly more current flows through the switching devices.

Other notes:

1. An interesting observation I made while experimenting with the SPICE model is that, once you've set the phase (i.e. time) relationship of the switches' "drain" voltage and current waveforms, this relationship is independent of the DC supply voltage. That is, as you raise or lower Vdc, you will not change where the drain voltage is, in time, with respect to the drain current. In other words, you shouldn't need to "retune" the resonant circuit as you raise or lower Vdc. This also implies that you can change the power out simply by raising or lowering Vdc.

(This observed independence on Power Supply level probably depends, at least in part, on how independent the value of C_ossis from the voltage across the Drain-to-Source of the MOSFET, which is something I don't know. Results might differ when using actual "real-world" components.)

2. Regarding the calculation of C3. After I finished my simulations I went back and reviewed some of the literature. There is a hint of how to determine C3 in the Cal Tech Thesis of Scott Kee (available as a PDF here). Refer to equation 7.18 and the discussion at the top of page 130.

The Thesis states, "it is apparent that the required fundamental frequency differential load is a resistance in parallel with an inductance having the same impedance at the fundamental frequency as the capacitance Cs does." Note that in our application Cs is the same as either C1 or C2 (C_oss).

To me this statement means that we can consider the transformer inductance to consist of two "virtual" parallel inductors, one inductor resonating with C3 at the operating frequency, and the other inductor resonating with C_oss, also at the operating frequency. In other words, the two inductors, as well as C3 and C_oss, can all be considered a parallel circuit consisting of these four components (all in parallel), resonant at the operating frequency.

The 400 nH inductance is actually the combined inductance of the our two "virtual" parallel inductors. Let's look at the circuit configuration during the half of the cycle when S2 is closed: C2 is shorted and both one end of the transformer's inductance (400 nH) and C3 are grounded. This means that the 400 nH inductance, C1 (500 pf), and C3 (3.655 nF) are all in parallel. The equivalent capacitance is therefore 4.155 nF, which, in conjunction with 400 nH, gives us a resonant frequency of 3.90 MHz.

3.90 MHz is pretty close to (although not exactly at) my operating frequency of 3.87 MHz. Coincidence? Should C3, when it's paralleled with C_oss, be calculated to resonate with the transformer's inductance at the operating frequency?

That is, should we use the equation:

F = 1/( 2 * π * √[ L(xfrmr) * (C3 + Coss) ] )

as a first-order approximation when deriving C3?

References:

Spice:

Free Download of LTSpice here.
LTSpice "Getting Started Guide" here.

Class E Amplifiers:

Sokal, "Class-E RF Power Amplifier", Jan/Feb 2001, QEX
Sokal, "Class-E High-Efficiency RF/Microwave Amplifiers", (a more detailed paper)
Lau, Chiu, Qin, Davis, Potter, Rutledge, "High-Efficiency Class-E Power Amplifiers" Part 1, May '97, QST, and Part 2, June '97 QST
Davis, Rutledge,"A Low-Cost Class-E Amplifier with Sine-Wave Drive"
Der-Stepanians, Rutledge, "10-MHz Class-E PowerAmplifiers"
Melia, Robert, O'Conner, Class-E Power Amplifier Design
Tayloe, Class E Amplifiers (Norcal QRP Presentation)
Class E Design Software from Tonne Software (I've never used this, so use at your own risk)
WA0ITP Class-E Amplifier Design Spreadsheet (I've never used this, so use at your own risk)
Class-E AM Forum
Class-E AM Transmitters (WA1QIX)
Class-E Amplifier Experiments , Calculations, and Notes on Designing Class-E RF Amplifiers, all by Bill Slade. (I've not verified the accuracy of these posts.)

Class E/F Amplifiers:

Taniguchi, Potter, Rutledge, "A 200 W Power Amplifier", Jan/Feb 2004, QEX
Letters, May/June 2004, QEX
Letters, July/August 2004, QEX
Kee, Aoki, Hajimiri, Rutledge, "The Class E/F Family of ZVS Amplifiers"
Kee, "The Class E/F family of Harmonic-Tuned Switching Power Amplifiers" (Cal Tech Thesis)
Kee, Aoki, Rutledge, "7.1 MHz, 1.1 KW Demonstration of the New E/F2,odd Switching Amplifier Class"
Jeon, "Design and Stability Analysis Techniques for Switching Mode Non-Linear Circuits: Power Amplifiers and Oscillators" (Cal Tech Thesis)
Kee, et al. U.S. Patent, No. 6,724,255
Niknejad, "Class E/F Amplifiers" (presentation, EECS 242, U.C. Berkeley)
Bohn, Kee, Hajimiri "Demonstration of a Switchless Class E/Fodd Dual-Band Power Amplifier (a 40 and 30 meter dual-band amp)

Improving the Heathkit HR-10B Receiver

2010-11-02T05:24:00.000-07:00

The Heathkit HR-10B is a 5-band, 7-tube amateur radio receiver manufactured from 1967 to 1975 and the companion to Heathkit's DX-60B transmitter. Essentially, the HR-10B design is the same as its predecessor, the HR-10 -- the only change seems to be that the top cover was painted with a wrinkly finish rather than the smooth finish of the original HR-10. It requires an external speaker or headphones.

I've always like the way the Heathkit HR-10 series receivers looked with their functional control layout and slide-rule dial. I picked this one up for a reasonable price, and I thought I'd give it a try. Powering it up, I immediately noticed a number of problems:

Slide Rule Dial tracking diverged greatly on 80 meters.
Receive frequency changed significantly as "RF Gain" was varied.
Couldn't use AGC (AVC) for SSB/CW modes.
Broad Selectivity.
Deaf on 15 and 10 meters.

From my internet research, these problems seemed to be common. Not a very good receiver, and it got me wondering...was there anything I could do to improve it?

Slide Rule Dial not tracking on 80 Meters.

If I calibrated the HF oscillator at the 3.5 MHz mark on the dial and then tuned up in frequency , I found that as I tuned towards 4 MHz, I would hear a 4 MHz signal at about the 3.96 MHz dial mark. In other words, as I tuned through the band the oscillator diverged significantly from the scale markings.

The tuning capacitor, when the dial is at 3.5 MHz, is at maximum capacitance. The fact that I'm receiving a 4 MHz signal at the 3.96 MHz dial tick tells me that the capacitance of the tuning capacitor has decreased too much as I rotated the dial.

One way to fix this is to add additional parallel capacitance to the 80-meter oscillator tank circuit so that , as the variable capacitor is tuned, the the overall "delta" in capacitance is reduced. I found that, for the amount of divergence that I was experiencing, paralleling C30/C66 with a 6 pf Silver Mica capacitor brought the dial into close enough calibration for my purposes. (There is a bit of divergence at around 3.6 MHz, but there's nothing I can do about that).

Receive Frequency shifts with changes in RF gain.

This seems to be a common problem with the HR-10 series receiver, and dynamic variation of the "125" plate-voltage line (i.e. the junction of R44, R43, and C56 in the schematic) seems to be the source of the problem. This voltage is generated by dropping the DC from the cathode of the rectifier through a series 1500 ohm, 10 watt resistor. Thus, because any change in RF gain changes plate current, plate voltage will also change because the the voltage drop through the 1500 ohm resistor has changed.

Unfortunately, as plate voltage varies, so does the frequency of the oscillator(s).

One way to fix the frequency shifting is to stabilize (regulate) the plate voltage. I added a series-string of four 5 watt Zener Diodes from the R44, R43, and C56 junction to ground (there's also a series 10-ohm resistor so that I can measure current, and thus power-dissipation, through the zeners). These diodes consist of three 33V, 5 watt diodes and one 18V diode, for a total voltage of 117 volts (prior to adding the zeners this node measured 144 VDC instead of the spec'd 125 volts, so there's headroom). Power dissipation in the 33V diodes measured to be about 1.2 watts apiece, so there's plenty of margin, dissipation-wise.

These diodes are shown at the top of the schematic below.

(Click on schematic to enlarge)

Note: Reference Designators in the schematic reference the original Heathkit parts. Parts without reference designators are new parts.

AGC (AVC) for SSB and CW

The HR-10B suffers from the standard problem with receivers designed pre-SSB: the AGC is worthless for SSB. Instead the user is advised to turn off the AVC, set the AF Gain to 3 o'clock (i.e. HIGH!), and then adjust the RF gain for an appropriate signal level.

In other words, there is no AGC for SSB or CW!

Before getting further into modifications, let's first take a step back and try to understand why this is...

Vintage receivers (prior to the days of product-detectors) typically used their diode-detector to detect both AM as well as SSB/CW detection. In SSB/CW mode this detector is driven by the output of a basic heterodyne mixer in which this output contains the BFO signal that it is driven with, as well as the beat products. These beat products form an envelope on the output waveform which is detected by the diode-detector. There are several problems with this method of demodulation for SSB and CW.

First, because the SSB or CW signal is demodulated with an envelope detector, the BFO signal must be quite a bit larger than the IF signal if there is to be minimal distortion on either CW or SSB. You can get an idea of why this is so by looking at the image below and comparing the envelopes of the two waveforms (Es = Eo and Es = 0.5Eo).

(Click on image to enlarge)

(Terman, F. E. Radio Engineers' Handbook, First Ed., McGraw-Hill Book Co., 1943, Page 567)

Imagine that Es is the IF signal representing a CW signal and Eo is the oscillator. If the amplitude of Es is significantly less than Eo, then the envelope on the resultant mixed waveform looks close to a sine-wave (look at the envelope of the Es = 0.5 Eo signal). And because this envelope is detected with the diode-detector, it will sound fairly undistorted.

But as the amplitude of Es approaches that of Eo, the envelope becomes much more distorted (look at the envelope of the Es = Eo signal), and thus the resultant detected output will be full of harmonics and sound grossly distorted.

So the IF signal must always be appreciably less than the BFO signal. But...this introduces another problem. Because AVC is also derived from the signal at the output of this mixer (which contains the BFO signal if its on), if the BFO is on this BFO component at the output of the mixer will swamp the AVC circuit and thus severely attenuate the receiver.

For this reason the operator manuals for older receivers state that, when receiving CW (or SSB) signals, the AVC should be turned OFF, the Audio Gain turned UP, and the RF Gain manually adjusted to provide a comfortable signal level. Not very convenient nor friendly to your ears when a very strong signal suddenly pops up nearby, and an excellent reason for adding a product detector and upgrading the AVC circuitry.

So...I decided to update the AGC circuit and at the same time add a product detector in lieu of the original CW detection scheme.

First thing I did was to replace V5, a triple-diode tube (6BJ7) with three 1N4148 diodes. (I had some DC voltage on the AVC line even with no input signal that I attributed to "leakage" in the 6BJ7 tube. Replacing the tube with diodes fixed this problem, and, of course, also lowered power dissipation).

For the AGC circuit I added a 1N4148 diode to change the AGC voltage-doubler configuration from a "Villard" circuit to a "Greinacher" circuit, which has better ripple characteristics. I increased the AGC decay time by paralleling a new 0.22 uF cap with the existing 0.05 uF cap (C29) and moving the location of the 1M resistor (R26) to increase the decay resistance from the original 2.2M ohms to 3.2M ohms.

With the 1M ohm resistor that had controlled the charge-rate moved, I replaced its function with a much smaller 9.1K ohm resistor (this value doesn't need to be exact -- in fact, you can probably get away with just using a jumper in lieu of this resistor).

I used an NE602 for the product detector -- my original goal being to use its on-chip oscillator for the BFO. Unfortunately, when I tried this (using the "stock" HR-10B BFO components) I found that the BFO frequency would "pull" with incoming signal strength (e.g. as RF Gain or AVC varied). I couldn't discover why this was happening, so I worked around it by replacing the on-chip oscillator function with a simple external oscillator using an MPF102 FET, and it worked much better.

If the BFO is on when in AM mode, you can hear it heterdyning with the carrier of the incoming signal, so it's necessary to turn the BFO off when receiving AM signals. To disable the oscillator the low-end of the oscillator tank circuit, T5, is removed from ground using a 2N7000 transistor. To turn the BFO on, this transistor must first be turned on to short pin 1 of T5 to ground.

While experimenting I ended up with quite a bit of attenuation at the input of the NE602 (the capacitive divider). I'm not sure if this much attenuation is needed; I added it because, during my testing I was experiencing some distortion issues and this seemed to help. However, I was making a number of changes around this time, and I could easily have over-compensated. Don't take these values as being the final word -- experiment!

The demodulated output from the NE602 drives two separate paths -- the audio path and the AGC path. I wanted to decouple the audio-path gain from the AGC-path gain (this is an audio-derived AGC circuit) just in case I needed different gains for the two paths. The op-amp inputs are fed via simple low-pass filters (to remove any residual RF from the output of the NE602). Gain of the audio path is about 37 dB, while gain of the AGC path is about 39 dB. This isn't much of a difference, and one could probably use the same op-amp to drive both paths.

The TL082 op-amp has a max power-supply rating of 35 volts (when powered with a single supply). I powered it with 30 volts to ensure that I'd have plenty of headroom when experimenting with gains -- the op-amps are biased at 15 volts, which give them about a +/- 12 volt swing (the TL082 output limits when within (roughly) 2-3 volts of either power-supply rail).

AGC gain is set to give me the same S-meter reading (roughly) when in either AM or SSB mode (BFO Off or On).

Audio gain is set to give the same audio ouput at the speaker (very roughly) when in either AM or SSB mode.

A relay is used to select between AM (no BFO) and SSB (BFO) modes. In AM mode, the HR-10B demodulation and AGC circuitry is the same as the "stock" receiver (with the exception of the changes to the voltage-doubler and RC time-constants described above). A 48V coil for the relay is used to minimize current drain (and thus power dissipation) -- it only draws 4 mA when on.

No changes were made to the Noise Limiter (ANL).

Here's a photo showing where and how I mounted the op-amps and the NE602. You can also see the string of zener diodes I added for oscillator stability near the top of the photo.

Note: Reference Designators in the schematic reference the original Heathkit parts. Parts without reference designators are new parts. And for many of these parts the value isn't critical -- I usually just pulled parts out of the junkbox that were in the ballpark of what I wanted.

Broad Selectivity

The HR-10B receiver has a two crystal crystal- lattice filter spec'd at 3 KHz down at 6 dB at an IF frequency of 1681 KHz.

Although 3 KHz might seem narrow, I've found that the skirts of the filter (on my receiver) are not very steep at all. This gentle roll-off of the filter skirts results in audio that is fairly broad, and, in fact, for AM reception I find that the receiver actually sounds pretty good.

I did find an article in Electric Radio regarding modification of the HR-10B crystal filter (as well as crystal filters in other receivers -- refer to the Electric Radio articles in the "Resource" section below). I decided not to attempt these mods at this time.

Deaf on 15 and 10 Meters

Lack of sensitivity on the high bands is a common complaint for this receiver, and mine is no different. I've poked around at this, and it looks like it's caused by a couple of things.

1. The RF Preamp (V1 and associated circuitry) has appreciably lower gain on 15 and 10 meters.

2. On 15 and 10 meters the HF Oscillator, rather than beating the incoming signal with the fundamental of the oscillator to get the IF frequency, instead beats the incoming signal with the second harmonic of the oscillator frequency. The amplitude of the second harmonic will always be less than that of the fundamental frequency, and, depending upon how the second harmonic is generated, the second harmonic might be significantly lower in amplitude.

Signal level at the output of the mixer is a function of the level of the input oscillator, so a lower-level oscillator signal will result in a lower-level output (and this can be exasperated if there's a square-law (or higher!) function in the mixing process.

I haven't yet looked into improving the performance on 15 and 10, given that there's sure to be a stability issue, too, given that we're using the second harmonic for the conversion. In other words, jitter or drift at the fundamental frequency means twice the jitter or drift at the second harmonic, and thus twice the degradation in stability!

Other Notes:

1. There's an optional Crystal Oscillator (HRA-10-1) which can be plugged in. This is a useful option!

2. Others have mentioned that alignment of the RF/Oscillator section can change when the bottom steel plate is reinstalled after completion of the alignment procedure. It has been recommend that holes be drilled in the bottom plate so that the receiver can be aligned with the plate in-place. I've not yet done this.

Resources:

Electric Radio Magazine Articles:

"Resurrection of a Heath HR-10B Receiver," Paschall, Issue 210, Nov. '06
"The Heathkit HR-10 Receiver," Hanlon, Issue 232, Sept. '08
"Heathkit HR-10 Receiver Update," Stock, Issue 234, Nov. '08
"Modifying Heathkit Crystal Filters," Stock, Issue 231, Aug. '08

Search Heath listserve archives here.

HR-10B Schematic here.

HR-10B Modifications here.

Standard Caveats

There might be mistakes. I cannot guarantee that everything is accurate. Use at your own risk!

The HR-10B has high voltages -- use caution whenever working on it!

WRL Duo-Bander 84

2010-10-26T09:00:00.001-07:00

I came upon a pair of these radios (plus an AC power supply) a few months ago while visiting a ham-radio store that specialized in older, used gear. It's a Duo-Bander 84, manufactured by WRL (World Radio Laboratories, Inc.) sometime in the late 60's.

(Click on image to enlarge)

The Duo-bander 84 is designed to be a sideband-only rig for 75 and 40 meters, and thus coverage on 75 meters is 3.8 to 4.0 MHz, and coverage on 40 is 7.1 to 7.3 MHz.

Tuning-up the transceiver's Transmitter is a single knob operation (rather than the usual peak-the-grid, dip-the-plate, and adjust-loading). Simply insert carrier into the Transmit signal by un-nulling the carrier balance (using the "NULL" control), and then adjust the "TUNE" knob for maximum output on the meter. (The "NULL" control (carrier-balance) will then need to be re-nulled).

The PA consists of a pair of 6HF5 sweep-tubes. With about 700 volts on the plates (the radio requires a separate power supply, by the way), I find that my peak power out is a bit more than 100 watts.

(A copy of the manual can be found via the link in the Resources section, below.)

Top view
(click on image to enlarge)

Bottom View
(click on image to enlarge)

Bottom view of a later version.

Note the new holes in the chassis under the IF board, and the crystal filter is sans case. I don't know if these changes were done in the factory, or were made later in the field by an end-user.

Notes:

1. Adjusting PA bias. Per the manual, one should set the PA bias so that, when the carrier is properly nulled and there is no voice-excitation (and the meter switch on the back panel is set to "Bias"), the meter needle, when transmitting, is on the "bias" calibration mark on the meter scale.

Unfortunately, one of my two radios had a broken meter. How should I then set its PA bias?

Well, I wasn't too sure how accurate the meter was on the rig that was operating properly (and note, the meters are not calibrated in mA), so I didn't want to use it as a calibration reference. I did a bit more research and discovered that the Swan 350 also used a pair of 6HF5 tubes in its PA. Their manual is a bit more explicit, and they state that the PA bias should be adjusted to 50 mA idle current when in Transmit mode.

I decided that if 50 mA is good enough for the Swan, it's probably good for the Duo-Bander, too! So I verified that the PA cathode resistance to ground was 2.5 ohms (there are four 10 ohm resistors in parallel), and adjusted the bias-pot on the back panel so that the PA cathode-to-ground voltage was around 0.125 volts.

And with the PA Idle Bias adjusted for 50 mA, the meter needle sits at the "Bias" marking on the meter faceplate (when the meter is in BIAS mode)!

2. No ALC circuit. The Duo-bander 84 does not have an ALC circuit to limit voice-peaks. To prevent excessive overdriving, I prefer, while monitor the output RF waveform with a 'scope, to adjust the mic's gain until the peaks are just at the peak-power out (this point will become evident as you adjust the mic-gain past this point -- the RF peaks will not get any higher, and you'll see more flat-topping).

I've thought about adding an ALC circuit, but decided that it wasn't worth the effort. For those who are interested in experimenting, a good place to start would be to look at the schematics for the Swan 350, the Galaxy V Mark II, as well as other radios (Heathkit HW-12A, Galaxy GT-550, etc.) that use sweep tubes in their finals. You'll see an ALC circuit that's common to all of these radios and which consists of a pair of diodes used as a negative peak detector to generate a negative ALC voltage based upon the PA grid voltage.

3. Distortion on transmit audio audio.

Both of my radios exhibited significant distortion on their transmit audio when I was first testing them. On both, I traced the cause back to a bad C9 capacitor. This is a 2 uF, 50V cap that acts as an AC ground for the collector-load of Q4. The balanced-modulator (Q6 and Q7) requires that the audio-drive to it (from Q4) consist of two signals 180 degrees out-of-phase and of equal amplitude. Thus the phase-splitter's (Q4) emitter and collector loads should be identical. If C9 is not a good AC ground for R11, then the amplitudes will not be equal, and there can also be a phase difference between the two that is not equal to 180 degrees.

On both radios I replaced their C9 caps with 4.7uF, 63V axial electrolytic caps that I had in my junk box, and the distortion problems greatly improved. (Note: it's OK to use a 4.7uF to replace the 2uF in this application. Larger value caps provide a "stiffer" AC ground for the audio signals, due to their lower impedance at audio frequencies).

Here's C9 in the schematic:

(Click on image to enlarge)

4. Excessive Transmit Audio Low Frequency Roll-off.

Per the alignment instructions, the filter passband can be "shifted" in frequency by adjusting the Carrier Crystal frequency with trimmer C39 (mounted next to the carrier crystal at the back of the radio). I found that, even with the crystal adjusted to shift the filter passband as close to the carrier as I could, I still had excessive roll-off in the low-frequency audio, so much so that it seemed as though the low-frequency cut-off was around 500 Hz.

Poking around the audio path with a 'scope, I discovered that there was excessive low-frequency roll-off occurring just after the mic-jack coupling capacitor C7 (0.01 uF). I paralleled C7 (0.01 uF) with a 0.1 uF cap (or you could simply replace C7 with a 0.1 uF cap), and this removed the excessive low-frequency roll-off.

Bandwidth is now 300 - 3000 Hz.

5. Carrier Crystal Oscillator stops oscillating in Transmit.

While I was trying to adjust the Carrier Crystal oscillator frequency to shift the filter passband so that the low-frequency cut-off was around 300 Hz rather than 500 Hz, I discovered that, as I rotated trimmer C39 towards one of its limits, the oscillator would stop oscillating when I was transmitting (but there was no problem in Receive mode).

As an experiment, I paralleled C15 (150 pf silver mica cap) with a 100 pf silver mica cap, and this seems to have cured the problem. My reasoning for adding this cap is: the oscillator would stop oscillating as its frequency was lowered, and so I assumed this meant that the trimmer cap was approaching its maximum capacitance. I decided to add capacitance across C15 (that is, to the fixed-cap side of the voltage divider formed by C15 and C39) in order to return the capacitance ratio between these two caps back to a value where the oscillator still oscillated. It seems to have worked, but I can't say that this is an optimal solution. Consider it a band-aid which fixed the problem for this particular transceiver.

(Note that you can easily add this capacitor by simply mounting it across the coax (from the Carrier oscillator) connected to the two pins at the back of the printed-circuit board, very close to the trimmer cap C39.)

6. Receive Distortion due to AGC:

While operating this radio I noticed that, for some signals, there was some subtle distortion on the receive audio. This problem seemed to manifest itself with stations who were using wide-band audio (e.g. lots of low frequencies). Fortunately, most signals I copied didn't seem to have this problem.

But the distortion was noticeable enough on a couple of stations with whom I talk regularly, and so I decided to look into it.

I noticed that, per the schematic, there are actually two AGC lines: an "AVC RF" line (to the grid of V6) and an "AVC IF" line (to the grids of V3 & V4).

The AVC IF signal has a very fast decay time constant (essentially, the decay of C21, a 0.02 uF cap, is controlled by R22, a 33K ohm resistor, which is C21's discharge path into C22, a much larger 0.22 uF cap).

AVC RF, on the other hand, has a much slower discharge -- C22's decay is controlled by R23 in series with R24.

(Click on image to enlarge)

Looking at the AGC (or AVC, if you prefer) signals with a 'scope, I noticed a potential problem with the AVC IF line. If you look at the top photo below, you'll notice there are a lot of "spikes" on its waveform (as the AGC goes more negative, there is more attenuation). These spikes are voice peaks at C21, and they quickly decay via R22.

One often sees this sort of AGC response in older receivers. I believe this fast-decay AGC is to limit the gain of fast transients (e.g. static crashes?) yet not have them affect the longer-delay AGC. Unfortunately, in my experience, these sorts of spikes can "modulate" the received signal and produce perceptible receive audio distortion, and often the receive audio will sound better if one can eliminate this sort of AGC modulation. (Of course, recognize that, in doing so, there can be a trade-off with limiting the gain of fast-transient noise).

For the WRL Duo-Bander 84, one way to clean up this distortion is to add more capacitance to C21 -- that is, make it larger. But rather than add another cap, one can simply short-out R22, the 33K ohm resistor between C22 and C21. This means that the AVC IF line's decay time constant becomes the same as the AVC RF line's time constant, and you can see the result on the AVC IF line in the lower photo, below.

(One issue with this sort of mod, though, in which the capacitance is greatly increased, is that the attack time will slow down (because you're charging more capacitance). The Duo-Bander's Receive AGC is audio-derived, and therefore, as with many other receivers with audio-derived AGC, you can get an audible attack "pop" at the start of strong signals. In theory, the mod I've made should exacerbate this type of popping, but I haven't noticed much difference, if any, in attack pops with or without this mod.)

The mod can be made easily without removing the PCB: To short-out R22, jumper the top of C22 to the top of R37, as shown in the photo below.

Again, I want to stress -- the original distortion that I was hearing was very subtle and only manifested itself on a couple of signals that I listened to regularly. For the most part, all other receive signals sounded fine. So this modification is certainly not necessary, and, if you do try it, it's quite possible that you won't notice any difference on the majority, if not all, of the signals you copy.

7. Other problems...

Here are some of the other problems I found in my two Duo-Bander radios:

Very little gain on receive. I traced this to R46 (47K, 1W plate-load for V7b) reading infinite ohms. Replaced with a 47K, 2W resistor from my junk-box.

Very low TX audio. I traced this to a faulty R9 (270K, 1/2 watt) in the Mic Preamp circuit (it read infinite ohms with an ohmmeter). Replaced with same value resistor.

8. Measuring resistance per the Manual's Resistance Chart. While trouble-shooting the rig I discovered that a number of resistances listed in the Resistance Chart were reading infinite ohms, despite their non-infinite values in the table.

These were typically plate resistors (or other resistors connected to the +400V B+ supply), as well as, for example, cathode-resistors in the TX path (e.g. V7 cathode). The reason why they were reading infinite with my ohmmeter was that they had no path to ground. (Note that the manual specifies its resistance tables in the chart to be "resistance to ground").

To make an accurate comparison of all resistances, per the chart, I'd recommend that you actually measure across the specified resistors, or you could ground the appropriate relay pins (e.g. pins 2 and 6 for Receive and Transmit B+, and pin 7 for TX cathodes) and then make your resistance measurements (in either case, though, be sure that all power is first disconnected from the radio -- remove the connector from the power supply to the Transceiver's P1 Power Plug!).

9. Power Supply:

The manual states that the power supply requirements are:

HV: 800VDC @ 400 mA
Low B+: 325/375 VDC @ 200 mA
Neg: -100 VDC @ 30 mA
12VDC @ 200 mA
12VAC (or DC) @ 5 A

Per one of the links in the Resource section below, three power supplies were available for the Duo-Bander 84:

Deluxe 400 Watt AC supply AC384A $89.95;
400 Watt DC Supply DC384A $99.95;
250 Watt AC Power supply AC48A $49.95.

In addition, other Galaxy power supplies allegedly can be used, too (I've not personally verified this, though), per this post on the Electric Radio Forum:

	12/05/02 08:47 PM \| 0 Good Guy Alerts \| WRL Galaxy Duo-Bander 84 power supply info wtd:
n3ibx New Member 1 Post 0 Good Guy Alerts Washington Crossing N3IBX Ignore User	Hello All, Would anyone know if the AC supply from a Galaxy 5 MkII will work with the Duo-Bander 84? Any assistance will be appreciated. Mod-U-Later, Joe Cro N3IBX Joseph Cro
	12/06/02 07:38 PM \| 0 Good Guy Alerts \| Galaxy Power Supply
Stu New Member 1 Post 0 Good Guy Alerts Elmira NY K4BOV Ignore User	Yes Joe, Any Galaxy power supply except the PSA-300 (came with the Galaxy 300) will run the Duo-bander without modification. The PSA-300 will certainly run a Duo-bander; but, the 12 pin Jones plug must be rewired to conform with the later Galaxy pin configuration. If you want to send me the serial number of your Duo-Bander, I may have some service bulletins for your particular series. Might want to give me the serial number of the power supply as well. They all work as I indicated; but, some are more capable than others. Stu/K4BOV

10. Some other radios using 6HF5 tubes in their finals:

Swan 350 and Swan 400
Drake 2NT
Galaxy 300, Galaxy V, and Galaxy V Mark II
Galaxy 2000 (Amplifier)
Hallicrafters HT-46 and SR-400

Resources:

1. Specifications and more information on the Duo-Bander 84 can be found here and here. (Admittedly these are both sketchy).

2. An Instruction Manual (in PDF format, and including alignment instructions and schematic) for the Duo-Bander 84 can be found here.

Standard Caveat:

Of course, I may have made a mistake, so use my suggestions at your own risk!

Also, this radio uses high-voltages that can kill you. Always use caution when working on a radio of this type.

813 AM Transmitter Accessory: External PTT Control

2010-08-21T10:19:00.000-07:00

My 813-based AM transmitter (see posts here, here, and here) sits about 8 feet behind me in the shack. I wanted to put the TX/RX antenna relay near my operating position (rather than at the transmitter which would have meant running an extra length of coax over to it), and I also wanted a conveniently-placed TRANSMIT switch next to my operating position so that I could easily put the transmitter into XMIT mode.

I also wanted to use my Flex-5000 transceiver as the receiver because it has, to my ears, a very good Synchronous AM detector and sounds great on AM. But I needed a way to "Mute" the 5000 via external control (sure, I could use my PC's mouse each time and click on the Console's "MUT" button, but what a pain -- I wanted automatic control). Unfortunately, the 5000 lacks an input that can be used as an external mute control for the receiver. But all was not lost ...

On the 5000's back panel there is an RCA connector for an external PTT input. I normally use this to place the 5000 into XMIT mode. But why not have it serve a dual purpose? That is, why not let it act as either a PTT control (its normal function) or a Receiver MUTE control?

I modified the Flex Console code so that the 5000's PTT RCA jack can be used for either of these two functions. And to select which function this RCA jack would perform, I added a new "button" to the Flex Console.

Let's start first with the circuitry. Here's the schematic:

(Click on image to enlarge)

Design notes:

Everything came out of my junkbox, which is why you'll see a 26 volt relay (the Antenna Relay) mixed with 12 V relays, and powered from a 24VAC transformer. I used what was at hand.
J7 connects to the PTT RCA on the 5000's back panel, and can either be used to place the 5000 into XMIT (via a pushbutton attached to J6), or as a 5000 RECEIVER MUTE control when I'm transmitting with the 813 AM transmitter.
J5 connects to the first set of contacts in the 813 Transmitter Sequencer (JP5 pins 1 & 2, per page 3 of the 813 AM Transmitter schematic). This keys the Antenna Relay as well as the 5000 Receiver Mute.
The military Antenna Relay internally shorts the unused RF port, as I've shown in the schematic.
Switch SW2 is a toggle switch and, when ON, it places the AM Transmitter into XMIT mode.
Originally relay K3 did not exist and switch SW2 connected directly to the AM Transmitter's PTT input (via jack J4). I quickly discovered, though, that if I toggled SW2 ON to transmit and if I'd forgotten to turn on power to my External PTT circuitry (that is, the circuit above), the Antenna Relay wouldn't switch-over during XMIT, and I'd transmit into a short. Not good! So relay K3 was added as a safety interlock. If power isn't ON, relay K3 is OFF, and switch SW2 cannot place the AM Transmitter into XMIT mode. SW2 can only act as a PTT switch if power is ON.

Here's the finished box (shown in Receive mode):

And here's a snapshot showing the new "button" I added to the Flex Console.

If this button is depressed (as it's shown above), the 5000's PTT RCA acts as a Receiver MUTE input. The software (modified by me) MUTES the 5000 and this button will turn RED whenever the PTT RCA is shorted -- that is, whenever the AM Transmitter is transmitting.

If this button is not depressed, the PTT RCA operates normally, that is, as an input for an external PTT control. In this mode, my external PTT push-button puts the Flex-5000 into transmit.

Caveats!

Standard warning applies: I may have made mistakes when writing this post or in my design. I cannot guarantee that everything is correct. Use at your own risk.

AM Transmitter, 813 Style, Part 3 (Everything Else!)

2010-08-04T05:12:00.002-07:00

This post (Part 3 in a three-part series) describes the 813 transmitter's power supplies, the control circuitry, and the meters. (Part 1 is here and Part 2 is here).

First, let's start with the High Voltage Supply schematic:

(Click on schematic to enlarge)

Notes regarding the above schematic:

1. The high-voltage supply is a standard full-wave rectifier into a capacitive-input filter. The caps are rated at 350 volts each, and so eight caps in series handle the high-voltage (overall the capacitance is 275 uF at 2800 volts). The high-voltage is equally divided across each cap with a 50K ohm resistor across each.

2. Additionally, there's a bleeder-resistor chain totaling roughly 40K ohms that can be connected across the entire capacitance bank to more quickly bleed off the high voltage whenever power is turned OFF. When the transmitter power is ON, this bleeder resistance is disconnected, so that power isn't wasted.

3. There is a 25 ohm power resistor to limit input-current surge into the transformer primary when the HV supply is first turned on. This resistor is then shorted-out when the HV reaches a certain voltage (refer to the Control circuitry schematic below).

4. A set of relay contacts is in series with the HV transformer primary. These contacts serve two purposes. First, when power is first turned (to the low-voltage supplies and to the filaments of the 813s), there's a delay of about 1 minute before the HV will turn on via these contacts. Second, if there's an over-current condition in either the PA or the Modulator, this relay opens to remove AC power from the HV supply. The STARTUP neon lamp signals when either of these two conditions is true (that is, when there's no AC across the HV transformer primary).

5. The HV transformer (I picked up in a trade) has multiple output taps and its input can be wired for either 120 or 240 VAC. I'm using its 3000V output taps and its input taps are wired for 240 VAC (but actually connected to 120 VAC). This gives me, with the capacitive-input filter, an idle plate voltage of about 2300 VDC.

6. Five 1KV diodes series-connected are used in each side of the full-wave rectifier. They don't have parallel resistors or caps to "equalize" their voltages because I'd read (some time ago), that modern diodes don't need this sort of protection. But I could be wrong -- you may want to add these.

7. Also, an unloaded HV of 2300 VDC implies that each diode string is seeing about 4600 VDC PRV (because the voltage across the capacitance string should be equal to the peak-voltage, given that it's a capacitive-input filter). This is getting quite close to the 5KV rating of the string; for a bit more margin I would recommend adding another diode in series on each side of the full-wave rectifier, so that each string has 6, not 5, diodes. (Under load (i.e. transmitting), the PRV seen by the diode strings drops.)

Here is the schematic for the Low Voltage Supplies:

(Click on schematic to enlarge)

This page of the schematics shows the PA grid-bias power supply, the PA screen power supply, the modulation transformer, and a voltage monitoring port.

Notes regarding the above schematic:

1. The PA grid-bias power supply uses series-pass regulation. When the transmitter is idle, this voltage is about -240 VDC. While transmitting, the zener-diode string voltage-reference is connected to the PA Power Return, and the grid-bias voltage goes to about -170 VDC.

Power resistors R93-R96 are used to share power dissipation with transistors Q6 and Q7 so that the transistors don't dissipate the brunt of the power when they are sourcing current. Depending upon the desired grid-bias voltage, different resistors in the R93 - R96 chain should be used (and the others shorted-out). For a grid-bias voltage of about -170 VDC, I've shorted-out R93 and R94. So, assuming a 25 mA total grid current, this means that about 39 volts will be dropped across R95 & R96 (about 1 watt, total dissipation) and 30 volts across Q6 and Q7. Assuming they share current equally, Q6 and Q7 will each dissipate about 0.4 watts.

2. There is a 40 VDC supply used for powering the Control circuitry as well as various relays.

3. The Screen-voltage supply is another bridge-rectifier circuit with capacitive-input filter. Its output voltage can be adjusted using the Variac (T3) connected to the primary of the transformer. Low-voltage windings of the transformer (T4) provide 7 VAC (roughly) to power various lamps (rather than loading-down the 40 VDC supply with their power requirements).

4. Screen-voltage can be switched between either 0 volts or 300-400 VDC with switch S10. Zero volts results in a lower power output from the PA and is useful when tuning the transmitter.

5. There are a pair of banana jacks on the front panel of the power supply so that internal voltages can be monitored with an external DVM or scope. Many of these voltages are scaled down to keep them under 50 VDC (for safety reasons). Voltage measurements are:

PA Control Grid Voltage (Tube 1) /10
PA Control Grid Voltage (Tube 2) /10
HV/100
PA Power Return (PA Plate current * 10 ohms)
Modulator Power Return (Mod Plate current * 10 ohms)
24 VDC
PA Screen Voltage /10

Here is the schematic for the Control Circuitry:

(Click on schematic to enlarge)

This page contains the Sequencer and its associated relays, as well as control circuitry to: 1. One-minute time-delay, HV Transformer input-current surge protection, and HV Power Fault detection.

Notes regarding the above schematic:

1. The sequencer is based upon a W2DRZ design and uses a bi-directional shift register to "nest" the relays such that the first one ON is the last one OFF. The clock-rate at which the sequencer marches from one relay to the next is controlled by the potentiometer R35.

2. U5D prevents the HV supply from being turned on for about 60 seconds after the low-voltage power supplies are turned on (e.g. the filaments for the 813s, whose filament transformers are on the same AC line as the low-voltage supplies) by not driving the HV_FAULT signal low during that time. (I don't know if one actually needs this type of protection for 813 tubes, but it was simple to add, and so I thought, "Why not?").

3. If either the PA or the Modulator plate-current exceeds 600 mA (over-current), either U5A or U5B will detect this and release their relay, which then turns OFF the relay whose contacts are in series with the HV transformer primary, removing 120VAC from this transformer. Also, the appropriate FAULT lamp (PA FAULT or MOD FAULT) illuminates when either of these fault conditions occur. If an over-current fault is triggered, this fault condition latches ON and 120 VAC to the HV transformer primary cannot be reconnected until the fault is first cleared by pressing the CLR FAULTS button on the front panel.

In other words, 120 VAC is only applied to the HV transformer if: 60-second-warmup-finished AND no-PA-overcurrent AND no-MOD-overcurrent. If any of these three conditions is false, then the relay that connects the primary of the HV transformer to 120 VAC will not turn ON.

4. When AC power is first applied to the HV transformer primary, surge current into the primary is limited by a 25 ohm power resistor in series with the primary. However, as the HV approaches its operating level, this resistor needs to be shorted-out so that there is no current-limiting into the transformer. U5C detects when the HV voltage reaches an appropriate point and drives a relay to short-out the resistor. (Currently, this trip-point is when the HV supply reaches 1000 VDC).

5. The /BLANKING BNC connector, J22, isn't used. I'd originally planned to connect a Tektronix 604 Display Monitor (essentially just an X-Y CRT) to use as a trapezoidal waveform monitor, and the /BLANKING signal would blank the CRT when in Receive mode, but I scrapped this idea when I discovered that I'd need to move my "Audio Sample" port (I describe this issue in more detail in the "Modulator Deck" post below).

6. Most of the lamps (except for the Fault lamps and the Startup lamp) are 6 volt lamps run from about 7VAC (series-resistors bring the voltage across the lamps down to about 6 volts). The 7 VAC comes from some low-voltage windings on the Screen-voltage transformer, which I use for the lamps so that I wouldn't unnecessarily burden the 40VDC supply with the lamps' current requirements. (The down-side of using these windings is that the brightness of these bulbs will vary depending upon the setting of the Variac used to set the Screen Voltage, but I'm willing to accept this compromise.)

The Fault lamps are 28 volt lamps because the signals that drive them are also used to "latch" the fault condition, and my latching design requires a DC "high" signal be fed back into the ULN2003A. I use the 40VDC supply for this, dropped-down appropriately with resistors (~28 VDC for the lamps, and ~20 VDC for the two inputs of the ULN2003A).

And here is the schematic for the meters:

(Click on schematic to enlarge)

The meters should be self-explanatory. I used whatever I had at hand that had the styling I wanted and that also had scales that did not need to be redrawn. For example, I used 0-50 mA meters to measure current in the 0-500 mA range.

The "Modulator Plate Current" meter can also be used to monitor RF Current (0-5 Amps, so that output power can be monitored). This feature is described more fully in the Addendum section below...

And finally, here's the Wiring Diagram showing the interconnection between the decks of the transmitter and with external equipment:

(Click on image to enlarge)

Here are some photos of the construction:

First, start with a THICK piece of sheet-metal and then...

...add some wood bracing.

Due to space constraints in the area designated for power supply circuitry, I mounted the HV bleeder resistors, AC surge protection resistor, and associated relays on top of the transformer:

(Heavy Metal!)

Building the PA Screen and Grid supplies, and what will be the sequencer.

The back plate, showing all of the I/O connectors.

Wiring it up!

With Modulation Transformer mounted.

After mounting the heavy bits, a homemade dolly helps moving it around the shack.

Meters, left to right: Mod Ip, PA Ig, PA Iscrn, PA Ip, HV

PA Grid Current, 0-50 Miles-Per-Hour!

The finished transmitter, in its rack and on-the-air!

Miscellaneous Notes on the Power Supply:

1. I wanted a Power Supply deck that I could slide in and out of the rack in case I needed to work on it, which is why I made the floor dolly and why I also added three handles to the power supply (one handle on the back of the plate to let me tilt the power supply deck from the dolly onto the bottom of the rack, and the two in front to let me push it into, or pull it out of, the rack).

Note: the two heavy transformers were installed after I'd put the deck on the dolly!

Addendum...RF Current Measurement

With the transmitter buttoned-up in its rack and the rack in the corner of the room, there was no good way to measure RF Output power during tune-up of the transmitter and Matchbox, given my shack configuration. After trying to tune up the transmitter a few times, I discovered I really wanted a meter right at the transmitter that I could watch while adjusting, say, the transmitter's LOADING control.

Because the Modulator Plate Current meter (the left-hand meter of the five) is of limited usefulness, I decided to make it a dual-function meter. That is, why not add a switch and some circuitry so that it could read either Modulator Plate Current (0-500 mA) or RF Output Current (0-5A)? Converting from RF Current to power is a simple conversion: assuming the tuner is tuned for an SWR near 1:1, then Pout = I*I*50. For example, a current of 2.24 amps equals 250 watts.

In my junk box I had an old Heathkit HM-2102 SWR meter from which I'd previously pulled out the meter (to replace a blown meter in an SB-220 amplifier: see previous posts) -- its RF box with its two SO-239 connectors would be perfect for the RF current transformer and rectification circuitry.

The circuit is straightforward and very similar to a design in recent ARRL handbooks (in their "Station Setup and Accessory Projects" chapter) -- see the schematic for the Meter Panel, above. I used 150 ohms in lieu of 50 ohms at the transformer's secondary so that I'd have enough voltage to drive the meter, and the 1:40 turns-ratio means that this load of 150 ohms is equivalent to an insertion of an additional 0.1 ohms in series with the RF line. In other words, loading effects are negligible.

Calibration was done against an LP-100 power meter at both 50 watts (1 A RF current) and 100 watts (1.41 A RF Current) into 50 ohms.

RF Current Sampler

2.3 Amps equals 264 watts.

How does it sound on the air?

You can listen to a clip of the 813 Transmitter on W6THW's website here. It's the track labeled "K6JCA 813 RIG (AM)".

(The rig was putting out about 300 watts, carrier power. Mic is a Heil PR-40 run through a Beringer 802 Mixer/EQ box, which feeds the Johnson Ranger's microphone input.)

References:

Sequencer Designs:

W2DRZ Sequencer

RF Current Sampler:

Refer to the High-Power Directional Coupler (that immediately follows the "Tandem Match" project) in recent ARRL Handbooks (e.g. page 22.42 in the 1997 edition of the ARRL Handbook).

Caveats!

Standard warnings apply:

First, I may have made mistakes when writing this post or in my design. I cannot guarantee everything is correct.

Second (and most importantly), this design uses high voltages that can kill you. Be cautious and BEWARE!

AM Transmitter, 813 Style, Part 2 (Modulator Deck)

2010-08-04T05:12:00.001-07:00

This post (Part 2 in a series of three parts) describes the modulator stage of my 813-based 75-meter AM transmitter. (Part 1 is here and Part 3 is here).

The modulator uses a pair of 813s wired as triodes and in a push-pull configuration. It's driven by an external audio driver (in this case, the audio from a Johnson Ranger). The modulation transformer is mounted on the Power Supply Deck, and can be found in the schematics posted in Part 3 of this design.

(Click on image to enlarge)

Here are some pictures.

Using a scrap chassis from the junkbox. Just a few extra holes!

Notes on the Modulator Deck

1. John Staples' "Electric Radio" article (issue #57, January, 1994) mentioned the use of a small amount of negative voltage (around -1.5 volts) as grid bias for the modulator tubes. I had originally designed a DC supply into the Modulator Deck to provide some small amount of negative voltage for biasing these grids, but, during testing, I discovered that it wasn't very "stiff" and would fluctuate significantly with voice peaks.

A quick test revealed, though, that 0 volts bias actually worked pretty well, and it had the added advantage that I could eliminate all of the bias supply components!

I left the ability to add a negative bias supply externally, though. An external bias supply can be connected between pins 6 and 7 of the Octal Jack on the back panel of the Modulator Chassis. If no external bias supply is used, then the bias must be set to 0 volts by shorting out these two pins.

(Note: with 1800 VDC plate voltage and 0 volts grid bias, idle plate current is about 23 mA through each 813 (connected as triodes), which equates to about 45 watts plate dissipation per tube.

2. I'm not sure if the suppressor grids of the 813s should be connected to ground or connected to the other two grids in the tube when using the 813's as triodes. I followed W6BM's example (and for which he plotted his curves) and connected all three grids together, as mentioned in his article and per his schematics which he kindly sent to me (these were not published in Electric Radio).

But I noticed that K1JJ (see the K7JEB website below), as well as W7XXX in his Electric Radio article (ER # 125), connected the suppressor grids to ground. The 1959 Edition of the Radio Handbook discusses "Zero Bias Tetrode Modulators" (section 30-8) and also shows a pair of 813's in push-pull with the suppressor grids grounded and the control grid and screen grid connected together. And W7XXX, in his Electric Radio article, alludes to potential stability issues (and the modulator becoming an unwelcome generator of RF), but unfortunately he doesn't provide any references.

(Digging further, I found a discussion on the AMFONE forum (here), and some mention of stability and hi-mu versus low-mu configurations, but again, nothing that I, as an engineer, would consider definitive.)

So -- is there anything wrong with connecting the suppressor grid to the other grids? Frankly, I don't know, and my research hasn't yet revealed any adequate explanations as to why this approach might be bad. But at least John, W6BM, plotted his tube characteristics using this connection configuration: so there is some data associated with it, and I decided to follow his approach. So far there hasn't been a problem...

3. When I first designed the Modulator Deck, I didn't have resistors in series with the grids of the two 813s, and I discovered during testing that modulator plate current would suddenly sky-rocket, tripping my over-current protection circuitry in the power supply. Adding 100 ohm resistors in series with the grids of each tube calmed them down. (Note: W6BM used 56 ohm resistors, but I didn't have this value in my junkbox, and 100 ohms seems to work fine.)

4. The 2K ohm resistors across each grid are supposed to provide a more constant load for the Ranger driving this Modulator Deck, per John, W6BM. John used 1.6K ohm resistors, but I had 2K's in the junkbox, so in they went instead. Are they really needed? I don't know -- a distortion test made with and without these resistors would answer that question.

5. I had originally added the transformer-coupled "audio sample" so that I could do trapezoidal monitoring of the transmitter's performance, and my thought was that sampling the audio prior to the modulator would be best, because then I could see if there were any non-linearities introduced by the modulator/mod-transformer. Well, it was a nice idea in concept, but it doesn't work in practice. Because of the time-delay through the tubes, you really must sample the audio at the output of the modulation transformer. Otherwise, you get a "phase distorted" trapezoid.

Old ARRL handbooks have photographs showing this type of distortion on the trapezoid waveform. For example, from the "Amplitude Modulation" chapter of the 1955 ARRL handbook:

Anyway -- rather than add another audio sampling circuit at the modulation transformer (and its high voltages), I decided that it would be sufficient to just monitor the RF itself using my already-existing "RF Sample" port on the PA Deck, and simply adjust the audio gain by look for "zero-lining" on the RF waveform -- after all, this was how I monitored the performance of my other AM transmitters, and it seems to work well. So my "Audio Sample" port on the Modulator Deck really isn't needed.

6. I added some 1 ohm resistors and test points (i.e. feed-thru caps) to allow me to measure the cathode current of each 813 independently, as well as overall grid current. Measuring each tube's cathode current allowed me to easily find a matched-pair of 813s.

7. For experimenting with grid-bias voltages, an external DC supply could be connected between pins 6 and 7 on the octal plug. For 0-volts grid bias, these two pins should be shorted together.

8. Although I grounded the bases of the 813 tubes in the PA Deck (using fingerstock), I did not ground the bases of the 813s in the modulator deck.

How does it sound on the air?

You can listen to a clip of the 813 Transmitter on W6THW's website here. It's the track labeled "K6JCA 813 RIG (AM)".

(The rig was putting out about 300 watts, carrier power. Mic is a Heil PR-40 run through a Beringer 802 Mixer/EQ box, which feeds the Johnson Ranger's microphone input.)

References

Articles:

"A Modern One Kilowatt AM Transmitter," W6BM, Electric Radio, #15, July, 1990
"813 Triodes as Modulators," W6BM, Electric Radio, #57, January, 1994
"Triple X 813 Homebrew Transmitter, Part Two," W7XXX, Electric Radio, #125, Sept., 1999
"Zero Bias Tetrode Modulators," Radio Handbook, 1959 Edition, Editors and Engineers, page 662
"Checking Transmitter Performance," ARRL Handbook, 1955 Edition, ARRL, pages 271-273

Websites, Modulator with 813s as Triodes:

AM Transmitter, 813 Style, Part 1 (PA Deck)

2010-06-01T16:07:00.000-07:00

(This is the first part of a three-part series. Parts 2 and 3 can be found here and here.)

[Note: I've changed the circuit slightly from my original publication in this post. Refer to the 19 August 2010 Addendum below.]

Some time ago I toured the shack of a friend, W7MS (Mike), in Reno, Nevada. I was very impressed by his collection of boatanchor equipment, but I was especially wowed by his RCA BTA-250M Broadcast Transmitter that he'd converted to 75 meter operation.

RCA's BTA-250M was designed to generate 250 watts carrier output using a pair of 813s modulated by another pair of 813s. Mike had done a great job of restoring his radio, and the four 813s, lit up side-by-side, were beautiful.

After I left I began thinking...I had a box full of 813s up in the attic. I wonder if...

Well, skipping ahead...about half a year later I finished constructing my 75 meter AM transmitter. Like the BTA-250M that inspired it, it too uses four 813 tubes: two in the PA and two in the modulator. It's designed to be driven by an external audio and RF source, and I use a Johnson Ranger to drive mine (identical to how John Staples, W6BM, drives his 813 transmitter, as described in Electric Radio, issue 15).

My transmitter generates output carrier RF power in the range of 200-350 watts. Here's the schematic of the PA Deck :

(Click on image to enlarge)

Notes on the schematic:

1. A large part of the design is based upon the 80-meter 813 amplifier described in "One-band Kilowatt Amplifiers," which can be found in the 1961 - 1968 editions of the ARRL Handbook. I designed a different pi-network using the equation in the Wingfield equations (reference recent ARRL Handbooks).

2. Per the original "One-band Kilowatt Amplifiers" article, the amplifier doesn't require any neutralization on 80 meters, so none was added.

3. There's a two-pole, three-throw rotary switch that's used to select screen-grid current monitoring (either the left tube, the right tube, or both tubes together). Monitoring screen current independently allows (allegedly) for tube matching.

4. Originally, I didn't have parasitic suppressors in the plate circuits of the 813s, but, when I first started testing the deck, I was seeing a lot of high-frequency stuff on the 'scope I'd connected to the "RF Sample" Output BNC, and I thought that this might be parasitics, so I added the two plate suppressors. They changed nothing, and I later discovered (using my spectrum analyzer) that the high frequency crud was all harmonically related to the fundamental -- that is, it's the natural byproduct of a Class-C amplifier, and that there were no parasitic oscillations.

I decided to leave the plate suppressors in (out of laziness), rather than remove them, but, per the "One-band Kilowatt Amplifiers" article, they shouldn't be needed on 80 meters.

5. Given the high-frequency harmonic components that I was seeing on my spectrum analyzer (up to and beyond 200 MHz), I built a metal cage around the entire PA deck to minimize unwanted EMI radiation.

6. The input network is the same as the one described in "One-band Kilowatt Amplifiers." C38 was changed from 0.001uF to 0.01uf to give a bit stiffer connection of the input network to ground (because there's no neutralization required, this capacitor doesn't need to remain 0.001uF that was used in the original article).

7. The pi-network's inductor is a three-inch long piece of air-inductor stock that I had in my junkbox (2.5" diameter, 6.7 tpi, 12 gauge wire). This length gives a max inductance of about 15 uH, but I tap it at around 10.8 uH.

8. To design the Pi-Network I first calculated the load that I needed to present to the plate using the equations for Class C RF Power Amplifiers found in the RF Vacuum Tube Amplifiers section of older editions of the "Radio Handbook," published by Editors and Engineers. (For my calculation I used 350 watts out (carrier) at a B+ level of 1650 VDC. This gave me a Plate Load (R_L) of about 2600 ohms.)

Then, given this load and the desired Q (Q should be in the range of 10 - 20; I chose 12), I used the Wingfield equations from the ARRL Handbook to calculate Pi-Network components. I put all of these equations into an Excel spreadsheet to allow easy manipulation and experimentation "on paper."

(Note: equation nomenclature changed in later editions of the Editors and Engineers "Radio Handbook" from that used in earlier editions, and I believe an error crept into the text. The best way to determine if an edition is in error is to compare the variable being solved-for in the description of the Class-C calculation steps (particularly steps 6 and 7) against the variable being solved for in the same steps of the "Sample Calculation" that follows this description. For example, the 18th edition of the book, the terms ebmin and epmin are swapped between their use in the description of the equations and their use in the "Sample Calculation" which follows the description. If you're putting your equations into something like Excel, watch out, or you'll have a problem! I fixed this by assuming the terms ebmin and epmin were correctly used in the "Sample Calculation," and so I swapped them instead when they were first mentioned in the prior description of the calculations.)

19 August 2010 ADDENDUM:

I noticed that, while transmitting, the RF power output would slowly increase from 250 watts (my initial setting) to 300 watts over a period of about 3 minutes of continuous transmitting. And this effect would reoccur after I let the transmitter idle for awhile (i.e. cooling down) and then began transmitting again.

In other words, it acted suspiciously as though the Pi-network's "loading" setting was changing with heat (its capacitance decreasing). So I pulled the RF Deck out of the rack for some bench testing.

If heating was an issue, as a quick test I transmitted for a few minutes, then powered-down (letting the HV decay to 0 volts!) and felt various components in the RF deck. Most felt OK, temperature-wise, but one capacitor, a 500 pf, 20 KV cap that I had placed in parallel with the "LOADING" variable-capacitor, was suspiciously warm. Hmmm...could this be the culprit?

I was using two sections (out of three) of the loading variable-cap (both sections connected in parallel). I wired in the third section of this cap (giving me 1800 pf max instead of 1200 pf) and removed the fixed 500 pf HV cap that was in parallel with the loading cap.

Powered back up, tuned the transmitter for 250 watts carrier output power, and after three minutes...it was still 250 watts! Problem fixed! Apparently the cap was lossy and, with the tank-circuit currents, it was heating-up and changing its capacitance.

I was a bit concerned that, with the three sections of the variable-cap wired in parallel, adjusting the LOADING control for a desired power might be a bit touchy because the capacitance might change too quickly as I turned the knob, but it's actually quite acceptable (admittedly, I have BIG KNOBS on my controls, which help when making fine adjustments). In this new circuit configuration, LOADING adjusts power from a min of about 180 watts to a max of about 370 watts RF output (carrier only, as measured using a Bird 50 ohm dummy load). I typically run the power at 300 watts carrier output.

The PA Deck schematic page (above) is now labeled "Rev. 2", to differentiate it from the original Rev. 1. The changes incorporated into Rev. 2, are:

Delete C81 (500pf, 20KV fixed cap).
Change C30 from a 1200 pf max variable cap to an 1800 pf max variable cap.

(Note: Other schematic pages are still at Rev. 1).

The culprit!

Fixed cap removed and 3 sections of variable-cap wired in parallel.

And here's the finished transmitter, up and running!

Some additional photographs showing construction of the PA Deck...

Building an RF "cage" around the PA using scrap sheet metal I purchased and had cut-to-size at a local metals recycling place.
I grounded the metal base of each 813 in the PA section at two different spots for each tube using flexible "fingerstock." (The fingerstock flexes out of the way whenever a tube is inserted or removed). I don't know if this is necessary, but I recall reading about it somewhere (can't recall where, though, at the moment).

The angled piece of black material between the tubes and the front panel is actually a rectangle of PCB material that I painted black and stuck into the PA Deck to deflect the fan's air up and out through the screen material on top of the case.

The finished PA Deck:

Other Notes:

1. The plate voltage when not transmitting is about 2300 volts DC, but it sags down to around 1800 at 250 watts out. This sag is probably due to the transformer itself coupled with the capacitor-input filter (rather than choke input filter) that I'd decided to go with (hey, I already had the caps in the junkbox). Modifying the supply to a choke-input filter may give me more output power, but honestly, it's more work than I think it's worth, so I'm leaving it as it is. (Note to self, though: next time, do a load test on the transformer and filter before installing everything!)

2. Screen voltage is about 350 volts idle and drops to 300 volts when transmitting.

3. At 290 watts out, HV reads about 1775 VDC, Plate current is 230 mA, Screen-grid current is 51 mA, and grid current about 20 mA. (Therefore efficiency is about 71 percent).

4. When running at a Pout of about 290 watts, plate voltage of 1775 VDC, ebmin of about 300 volts (assumed), and efficiency of 70% (plus other assumptions per the "Radio Handbook" equations) these numbers work out to a plate load of about 4000 ohms. For an inductance of 10.8 uH in the pi-network, Q (given a 50 ohm load) calculates to be about 17, so we're in the ballpark of a Q between 10 and 20.

5. Note the following pi-network Q relationships (using the Wingfield equations):

As output power increases (by changing loading capacitance), for a given value of pi-network inductance, pi-network Q will decrease.
As frequency decreases, for a given value of pi-network inductance, pi-network Q will increase.

"A Modern One Kilowatt AM Transmitter," W6BM, Electric Radio, #15, July, 1990
"813 Triodes as Modulators," W6BM, Electric Radio, #57, January, 1994
"An AM Kilowatt Using 813s 1989 Style," WA4KCY. Electric Radio, #5, September, 1989
"One-band Kilowatt Amplifiers," ARRL Handbook, 1961 - 1968 Editions, ARRL
"Class-C Amplifier Calculations," Radio Handbook, Editors and Engineers, 18th Edition (1970) [See note in text above re: error in equations.] Or one could use an earlier edition of this book, such at the 15th edition (pages 153-156) which doesn't have this error.
"Tank Output Circuits,"ARRL Handbook, 1997 edition, ARRL, pages 13.5 - 13.9 (Describes the Wingfield pi-network equations.)

Websites, Transmitters

Websites, RCA BTA-250M Manual

BTA-250M Manual

Websites, 813 Data

Detailed RCA data here or here.
813 Data, AA8V

Sequencer Designs:

W2DRZ Sequencer

Hooking Up a Western Electric 211 "Spacesaver" Telephone

2010-02-17T12:48:00.000-08:00

I picked up this cute little telephone recently at the local De Anza swapmeet.

From web research, I discovered it's a Western Electric 211 Telephone (their "Spacesaver" model). Typically you'll see these with dials mounted on top of the metal box (they actually look pretty cool). This one is sans dial, making it a "manual" (rather than "dial") telephone.

This set consists of a G1 case and an F1 handset. When I opened it up I discovered that the case only contains a hookswitch and nothing else! The drawing below shows a "generic" 211's circuitry (the simpler wiring of my "no dialer" telephone is shown in yellow). Note that the circuitry at the left is contained in a separate "subset" unit (which I will discuss further, below).

(Click on image to enlarge)

As I mentioned above, these telephones are meant to be used with an external "Subset" (such as the Western Electric 634 or 684), which contains a network for the anti-sidetone circuit as well as the ringer bell.

To test the telephone without a subset, I simply connected one side of the phone line to the "GN" terminal, and the other side of the phone line to the "Y" terminal. This will place the hookswitch, the mic element, and the receiver element all in series and connected to the Telco line. This implementation is shown in the photo below.

But there are several drawbacks when wiring the set this way. First, there's no anti-sidetone circuit, so you'll sound fairly loud to yourself in the receiver when you're talking into the mic. Also, there's a DC voltage across the receiver element, and thus there's the possibility (allegedly) that, over time, this might demagnetize it. Finally, there's no suppression of "clicks" in the receiver when going On or Off hook -- if you're holding the handset to your ear while "flashing" the telephone hookswitch, you'll hear VERY loud clicks.

So it seemed that the thing to do would be to find an appropriate subset. I looked around for one, such as the 634A, 684, or the 685A, but all the ones I found were fairly expensive, and I really didn't want to spend a lot of money on one.

Hmmm...could I build my own?

Well, I didn't need a ringer because I have a phone with a ringer across the room on my desk. I really only needed a network. If I could find a network, I could build my own subset...

One network I found was a "101A" network that was used in telephones such as the Western Electric 302 . This is just a transformer, and it requires an additional 2 uF cap to work with the telephone. But the 101A is a bit primitive -- it doesn't include any components to limit clicks. I decided instead to use a more modern network such as the 4228 or the 425 (the 4228 is found in Princess phones (I believe), while the 425 (e.g. 425E) is found in the 685A subset and 500-series telephones) . In the end I purchased a 4228 network from someone I found via the "Telephone Collectors International" List Server (on Yahoo).

Here's how I connected my 211 to my external 4228 network:

(Click on image to enlarge)

I want to stress, these connections are only for my "no-dial" 211 set! Here's a list of instructions for making these connections:

Required:

4228 (or 425) network
5-wire cable
211 (sans dial) Telephone

Instructions:

At the 211 Telephone, move the Handset's BLACK wire from the "BK" terminal to one of the two unused terminals marked "B".
Using the 5-wire cable, connect one end of one of its wires to the "L2" terminal on the 4228 network, and connect the other end of this same wire to the "Y" terminal in the 211.
Using the 5-wire cable, connect one end of one of its wires to the "GN" terminal on the 4228 network, and connect the other end of this same wire to the "GN" terminal in the 211.
Using the 5-wire cable, connect one end of one of its wires to the "B" terminal on the 4228 network, and connect the other end of this same wire to the "B" terminal in the 211.
Using the 5-wire cable, connect one end of one of its wires to the "C" terminal on the 4228 network, and connect the other end of this same wire to the "BK" terminal in the 211.
Using the 5-wire cable, connect one end of its last wire to the "R" terminal on the 4228 network, and connect the other end of this same wire to the "R" terminal in the 211.
Connect one of the two Telco wires to the "RR" terminal on the 4228 network.
Connect the other Telco wire to the "L2" terminal on the 4228 network.

Here's a picture of my homebrew "subset" with the 4228 stuck to the bottom of a project-box with hot-melt glue. Telco comes in via an RJ connector on the left side. The gray wire bundle (5 wires) goes to the 211 set...

(Click on image to enlarge)

(There's a drawing HERE to help you identify the terminals on a 4228 Network. Note that "L2" is a dummy terminal (that is, it isn't connected to anything), and you can use its two screws to connect one side of the telco tip/ring pair to the telephone, as I've done.)

Here's the phone, mounted for use by the workbench!

Resources:

1. Information on the 211 (and other old Western Electric phones) HERE.

2. Wiring & other info for the 211 (towards the middle of the post) HERE.

3. A source for old phones and parts (and subsets!) HERE.

4. 4228 Network information HERE and HERE.

5. "Telephone Collectors International" webpage and list-server.

Standard Caveat:

And of course, there's the standard caveat! I may have made a mistake, so use at your own risk!

Improving the Selectivity of the R-105A Receiver

2010-01-04T16:16:00.001-08:00

I discussed my experiences getting my ARR-15/R-105A receiver on the air in a previous blog posting (here). As I mentioned in that posting, the selectivity is very broad -- so broad that, when using the receiver on, say, an 80 meter AM net, it really suffers from adjacent SSB interference.

I would like to eventually pair this receiver with an ART-13 transmitter I'm working on, but first I needed to improve its selectivity.

How best to do this?

The R-105A has a variable IF -- that is, the IF frequency varies from 450 KHz to 550 KHz, with it being 500 KHz at the BFO "detent." Interestingly, when in MCW mode (i.e. AM), the BFO is OFF when the BFO dial is in its detent ('0') position, but it turns ON as soon as this dial is turned away from its detent.

So, clearly, for AM operation the BFO dial should be left in its detent position (otherwise the BFO is ON and you'll hear it hetrodyning with the carrier).

OK -- so the IF frequency is 500 KHz when copying AM. I wondered if I could cobble a 500 KHz AM mechanical filter into the radio. But first, I needed to find a filter...

I looked around the internet to see if I could find one for sale. Finally I found one on the Fair Radio site (Lima, Ohio). Actually, I found two:

The first filter was contained within the AF Audio Amplifier Module (Fair Radio p/n 546-6053) for the Collins 618T HF Transceiver, and it was a 6 KHz wide, 500 KHz mechanical filter (F500 Y60 (526-9378)). The price Fair Radio was asking for this module, including filter, was $52.00.

The second filter was contained within the 500 KHz IF Audio Amplifier module (Fair Radio p/n 105-AA) for the RT712 / ARC-105 HF Aircraft Transceiver. Fair Radio doesn't specify the filter model number, except to say that it's an AM filter, and that the IF frequency is 500 KHz. The price for this module was $44.00.

(Fair Radio provided the latter with a copy of the module's schematic. I don't know if they also provide this if you order the first module.)

Fair Radio also mentioned that the ARC-105 is a pressurized version of the 618T2 transceiver, so I thought, what the heck, it probably has the same filter as the 618T series transceivers, which is the F500 Y60. And best of all, it was $8 less than the other module.

So I ordered it. When it arrived, I noticed that all of the boards within the module were covered with a clear conformal coating. This wasn't a real big deal, but it's probably one reason why this module costs less than the 618T module. And it kept me from determining what the filter part number actually is, because the conformal coating effectively glued the shield bracket to the filter, and the filter's label is under it.

Here's the module I received from Fair Radio (with its cover removed). The filter is the long cylinder in the upper corner...

Because of the filter's insertion lose, I would need some sort of amplification to compensate. I first tried using the amplifier built onto the filter module's PCB. It actually worked fine, and I initially considered using this PCB with filter and amplifier already built and working for my project, but the board itself is a bit too large for mounting within the R-105A chassis, and so I decided to roll my own amplifier.

I designed and bread-boarded the filter and amplifier circuit and, after some changes, I mounted them on a small piece of copper-clad PCB material. Here's the finished circuit, installed within my R-105A:

This design uses an existing hole within the receiver for mounting, so a purist can, at a later date, easily remove my modification and return the receiver to its original condition. I connected this circuit into the receiver's existing circuitry by unsoldering a wire from one of the pins of the IF transformer Z-119 and soldering in new wires in its place (and connecting another wire to the receiver's AVC line). Again, the new wires can be easily removed and the original wire reinstalled to return the receiver back to its original condition.

Here's a comparison of the receiver's bandwidth with the filter in and out (I've raised the filter's trace by 10 dB so that you can more easily see the difference).

(The above measurement was made by driving the receiver's antenna input with a wideband noise source (General Radio 1383 Random Noise Generator), and then, using a FET probe attached to the grid of the first IF amplifier, capturing the frequency response on my HP 8568B spectrum analyzer).

Here's the schematic:

(Click on image to enlarge)

Notes:

1. I don't have a data sheet for the filter. I'm leaving its drive impedance high (it's the first mixer's load, which, when the filter is switched-in, is simply receiver's IF transformer, unloaded). And I've set the filter's load impedance to 2K ohms. Correct? I've no idea.

[Update, 10 Jan 2010: I just came across the following schematic in Service Bulletin 2 for the Collins 51S-1 receiver in which they install a F500 Y60 filter into the 51S-1. Note that the newly-installed AM filter (bottom half of page) uses a total of102 pF (51 + 51) at the input of the filter and 113 pF at the output (51 + 62). Source impedance is a 10 mH inductor (coupled to filter via 1000 pf) and load impedance is 220K ohms (coupled via 470 pF). I've experimented with using a 10 mH inductor that's switched-in (via the relay) to be the load, in lieu of using the existing transformer (as I'm doing now), and I've also experimented with different values of filter input and output capacitance. When I varied the capacitance I really couldn't see any significant change in filter shape/symmetry, and different 1sy Mixer loads produced different gains, but...nothing else of any real significance, so, at this time, I don't plan to change my design. But I'm including the Collins schematic below in case you'd like to experiment, using it as a basis for your design...]

(Click on image to enlarge.)

2. I installed variable caps (and some additional capacitance using silver-mica caps, similar to what is used at the filter input and output on the RT-712 module) at the filter input and output of my circuit, but during my testing I couldn't really find much difference in filter shape and symmetry with the caps installed or not installed. So I removed the additional silver-mica caps, but left the variable caps (which were peaked to give max response). These could probably be removed, too. The schematic reflects the current implementation that uses only the variable caps.

3. Because of the conformal coating on the PCB, you need to remove the filter and its shield bracket together. But this isn't too difficult. I simply clipped the bracket's four mounting tabs flush to the PCB with a pair of diagonal cutters. The stubs of the tabs that remained on the bracket were sufficiently long enough to allow me to easily solder them to my copper-clad board to hold the filter in place.

4. The relay came from the RT-712 module, too.

5. The filter is switched into the circuit when when the receiver's front panel Channel switch is switched to Channel 10, and only Channel 10. For all other channels the filter is out-of-circuit, and the receiver's selectivity is back to its original bandwidth. (Note: The filter could be switched-in for more channels -- simply connect the additional channels at the Channel switch using individual diodes in a wired-OR wiring.

6. The pot is set to give the same AVC voltage, for a given signal, when the filter is in-circuit, compared to when it is out-of-circuit.

7. [Note (7 January 10): I changed the design slightly (raising the source impedance for the source feeding the filter, lowering the gain of the first transistor stage) from when I first published this post, so the photo above of the implementation doesn't exactly match the schematic, and the schematic revision is Rev B, not Rev A. The following discussion pertains to this new revision]

I first designed the amplifier with only one transistor, but when I tested it I found that there wasn't quite enough gain, so I added the second transistor. I didn't rebalance the gains when I added the second transistor, so the first transistor supplies the majority of the gain (41 dB, assuming an Ic of 0.9 mA and a load of 3.9K || 100K || 33K (Av = gm*Rl)). The second transistor is variable gain, and in my case the pot is set to 3.8 Kohms, so the gain of this second stage is about 9 dB (Av = Rl / Rf). Therefore, the overall gain calculates to be about 50 dB.

50 dB is a lot of gain. I don't know why it needs so much. Hmmm...could I have made a math error in my gain calculation above?

Nope. I just simulated the amplifier using LTSpice IV (a great program, available here for free), and at 500 KHz the gain is 49.3 dB in the simulation.

Why so much loss? I don't know, but...I don't plan to investigate any further: the mod works to my satisfaction, and I've other projects to work on!

8. You can find schematics for the 618T series HF transceivers here.

9. And, as a reminder, my earlier post on the R-105A is here.

And of course, the following always applies...

Standard Caveat...

I hope you find this information useful, but please, use these modifications at your own risk -- although they worked for me, I cannot guarantee that they'll work for you. (After all, I could have made a mistake in transposing them from my lab notebook to this post.)

If you do find any errors, or if you have any questions, please let me know. Thanks!

- Jeff, K6JCA

Revisiting the Heathkit Cheyenne Transmitter

2010-01-02T08:51:00.000-08:00

[Note, 24 Feb 10: I've just updated my schematic to fix an error. I'd forgotten to break the original connection from V6 pin 5 to ground. That's now been fixed, and I've updated the schematic revision to Rev. 2. I've also added Notes 5 & 6 at the end of this post. - Jeff]

(Cheyenne, now with knob and case. HP-20 power supply by the side)

In my earlier Cheyenne posting (click here) I described the modifications I'd made to the Cheyenne's audio stage to improve audio fidelity. This new post will describe some additional changes I've made since that original post, and I'll also update the MT-1 schematic.

I had been noticing that, during transmit, the 6CL6 tube (driven by the 6AU6 VFO) was getting very hot. I temporarily put a 0.5 ohm resistor in its plate circuit so that I could measure its plate current, and I discovered that it was drawing on the order of 64 mA, which, given its plate voltage of 322 volts, was 21 watts of plate dissipation.

The 6CL6 has a maximum plate dissipation of 7.5 watts, so this was far beyond where it should operate.

I took a look under the chassis and immediately noticed that the 6CL6's screen-grid resistor was 6.8K, rather than the 27K shown on my Heathkit schematic. But, when I referred to the assembly manual for verification, the instruction for the installation of this resistor describes a 6.8K resistor. In other words, the assembly manual's instructions match the wiring of my transmitter, whereas the schematic does not match the wiring.

I decided that the easiest way to lower the 6CL6's plate current would be to lower its screen-grid voltage. To do this, I added 47K ohms in series with the existing 6.8K ohm resistor. This drops the plate current to about 23 mA on 80 meters (7.4 watts plate dissipation) and 15 mA on 40 meters (4.9 watts).

Also, I found that I needed to change the value of the screen-grid resistor to the 6146 PA. I had initially changed this to 50K (please refer to my first post), but, after lowering the 6CL6 plate current (and replacing the 6146 tube) I found I needed to double this value to 100K ohms to give me my desired "sweet spot" of 10 watts out when the pot has been adjusted so that there's about 6 volts on the cathode of the first stage of the 6DE7 modulator (V6 pin 5).

While making these changes I also discovered other differences between the wiring of the Cheyenne's RF stages and the schematic. Again, the wiring within my Cheyenne corresponds to the instructions in my assembly manual, rather than the schematic

So I've modified the schematic to include my mods as well as the differences that I've found (to date) between it and the actual transmitter wiring (as described in my assembly manual). My mods are in red, and the wiring differences are in green:

(Click on Image to Enlarge)

These differences between the schematic and the manual are:

V2 screen-grid resistor is actually 6.8K, not 27K.
V3 screen-grid resistor is actually 4.7K, not 27K.
The tuning circuit for the PA grid is not a parallel L-C circuit, but instead it is a pi-network. 300 volts DC is fed to the plate of V3 (5763) via a 2.5 mH inductor. The signal from this plate is then coupled to the input of the pi-network via a 6.8 pF cap, and the output of the pi-network's inductor is directly connected to the PA grid (V4 pin 5), to which a grounded 47 pF cap is also attached.

There may be other differences, too, but I've not yet come across them. And unfortunately, I don't know which set of changes are the later (and thus, I assume, better) changes.

Additional Notes:

1. I used a Krylon spray-paint that I found at the local hardware store to repaint the Cheyenne's cabinet. It's Krylon Indoor/Outdoor Gloss Hunter Green. This isn't an exact match for the original Heathkit paint (which is slightly lighter and, in my opinion, contains a bit more blue), but I find it is close enough for my tastes.

2. With a new 6146W and the 6CL6 mods, for 10 watts out on 80 meters, the 6146 Screen Grid voltage is now about 94 volts (compared to the (roughly) 52 volts I describe in my original Cheyenne posting (for 9 watts out)).

3. There is a change to my tuning instructions in my original post: Now, when tuning, I peak the Drive control (grid) to give me maximum power out. Note that there may be two peaks (one lower than the other) as you rotate the Drive control. On 80 meters I get max power out with a grid current of about 2 mA, which is not the max grid current that I can get by rotating the Drive control.

4. The audio, in my opinion, sounds quite nice. I'm using a Heil PR-40 mic that is equalized using a Behringer 802 mixer/equalizer (the PR-40 the highs and mids need to be accentuated), and the Behringer level is set such that the Cheyenne's Audio pot is at about 2 to 2.5 on its scale. The Cheyenne's RF output is then fed into an Ameritron AL-811 linear.

5. If your power supply supplies a different high-voltage than my HP-20 (660 VDC), you may need to make some component value changes. For example, if you find that the final resistance of the 25K pot (from V6.5 to ground), after adjustment, is very small, you should increase the capacitance value of the 1uF cap that parallels it, otherwise you could lose low-frequency response in your audio. Or, if you want to keep this pot at roughly mid-level, you may want to try lowering the resistance of the screen-grid resistor to give you more output power.

My recommendation would be that you adjust the pot to give you (very roughly) 6 volts from V6.5 to ground, and then, if you don't have adequate power out, adjust the PA screen-grid resistor value. (If the voltage at V6.5 is too small, you run the risk of clipping the positive peaks of the audio at the grid of V6, which is why I recommend that the voltage at the cathode be about 6 volts).

6. By the way -- placing the 25K pot in the first section of V6, rather than the second section of V6 ( where others have placed it when modifying DX-60's), means you don't need to have a high-power pot (which you do if you place this pot in the cathode-circuit of the other half of V6). Instead, with it connected to V6.5, this pot adjusts the operating point of the first half of V6 (thus there's a low voltage across it, about 6 volts). Because the plate of this first half of V6 is dc-coupled to the grid of the second half of the tube, changing the cathode-voltage at V6.5 also changes the cathode voltage of the second section of V6, and thus changes the screen voltage to the PA.

(Raising the voltage at the V6.5 cathode drives this section towards cut-off (lowering plate current), which raises the voltage at the corresponding plate, V6.6, and thus the voltage to the grid of the second section. Raising this grid voltage means more current will flow through the second section of V6, which raises the voltage at the second cathode and thus raises the PA screen voltage.)

7. And as a reminder, my earlier posting on the Cheyenne is here.

8. An audio clip of my Cheyenne driving an Ameritron AL-811 amplifier can be found here.

Standard Caveat...

I hope you find this information useful, but please, use these modifications at your own risk -- although they worked for me, I cannot guarantee that they'll work for you. (After all, I could have made a mistake in transposing them from my lab notebook to this post, or there may be other problems in my rig that make its performance different from yours. So always verify for yourself that the changes have improved, rather than worsened, performance!)

If you do find any errors, or if you have any questions, please let me know. Thanks!

- Jeff, K6JCA

How to Make Your Equipment Look Like a Million Bucks!

2009-12-19T14:05:00.001-08:00

There's a lot of surplus equipment out there that can be used as the basis for a homebrew project. Here's a way to make it look more professional.

For example, let's take this...

...and make it look like this:

It's actually very easy. Here are the steps I follow.

First, For the panel, I measure its dimensions and the locations of all holes I want to use. Then, using an electronic drawing program (I use Adobe Illustrator, but others I know use Autocad), I accurately place all hole locations as well as any labels I want to add on my panel drawing.

I then print out the panel drawing. If the panel is larger than, say, 8.5 x 11 inches, you'll need to create a "B" size sheet (11" x 17"). My printer does not print B-size sheets, so I instead print two "A" size sheets, trim their edges, align them (note the alignment marks in the overlap area), and then tape them together so that they form a larger sheet. If you don't have a light-table for accurately aligning them, a window works fine...

I then go to the local copier store (here it's the "Kinko's" chain) where I copy my "composite" drawing onto a B-size sheet (so that it's a single piece of paper, not two taped together).

(If you can squeeze in an extra panel, do it, just in case you "goof.")

I then laminate it (the local Kinko's has a laminator, and it's free!)

And I cut out the panel overlay (and any large holes) with an Xacto knife...

Finally, I glue my overlay to the original metal panel using "Adhesive 77" spray adhesive (manufactured by 3M) and cut out the remaining holes, using the holes in the original panel to guide my Xacto knife.

And voila!

[Many thanks to Dick, W1QG, who taught me this technique. His great looking panels on the equipment he built were an inspiration.]

Converting an HP Counter into a Nixie Tube Clock

2009-12-14T15:47:00.001-08:00

Hmmm...I had an old HP 5233L nixie tube counter with 6 nixie tube digits gathering dust in the corner. What to do with it...?

Let's see, six digits...why not make it into a clock? After all, nixie tube clocks are pretty cool. And the counter came integrated with almost everything I needed: power supply and even a case with nicely milled holes. Mechanical work would be minimal, which defines my ideal project!

And as a nice bonus, there was also a back-panel BNC input for an external frequency reference, so I could use my GPS-locked frequency standard to keep the clock's time from "drifting" over time.

Conceptually, I figured it should look something like this (this includes a simple method for setting time):

Block Diagram

(Click on image to enlarge)

Pretty straight forward. But there was a small complication: I didn't have schematics for the 5233L.

This really didn't prove to be much of a problem. HP used the same nixie "single-digit" decimal counter plug-in module in a variety of its counter products. And I did have a manual for an HP 5232A. A quick glance revealed that it had the all-important schematic for the decimal counter module (HP part number 5212A-4A -- and although my counter used 5212L-4A modules, the 5212L-4A design seemed to be very close (if not identical) to that of the 5212A-4A)).

That manual, plus some scope probing of various signals within my 5233L counter, told me pretty much all I needed to know.

Some things I needed to do were:

Change two of the counter modules to count from 0 to 5 (instead of 0 to 9), so that minutes and seconds would each count from 0 to 59, instead of 0 to 99.
Add a reset circuit to the two "hours" digits such that, if the hour count increments to '13', it immediately resets the two hour digits.
Modify the Hour LS (Least Significant) digit to reset to '1' instead of '0' (so that hours count from '1' to '12'.
Disable the "Storage" feature of the Decimal Counter modules so that we can see the module counting.

So first, to count from 0 to 5, the 5212L-4A assembly(or 5212A-4a, or other variants) can be modified as follows:

Remove R45, R50, and C10
Change C11 from 200 to 470 pF
Move R59 from CR12 to CR11

If you're trying to understand how the decimal counter circuit works from its schematic, note that it counts per the following pattern:

D C B A
0 0 0 0 (0)
0 0 0 1 (1)
0 0 1 0 (2)
0 0 1 1 (3)
0 1 1 0 (4)
0 1 1 1 (5)
1 1 0 0 (6)
1 1 0 1 (7)
1 1 1 0 (8)
1 1 1 1 (9)

So, to count from 0 to 5 instead of from 0 to 9, we can use the transition of B from 1 to 0 to set C to 0. And D must be always kept at 0.

(Important schematic note: Q1/Q2 control bit A, Q3/Q4 control bit B, Q5/Q6 control bit D (not C!), and Q7/Q8 control bit C (not D!)).

For item 2, the Hours Reset circuit needs a few more parts. I mounted them under the counter chassis.

(Click on image to enlarge)

For item 3 -- to modify the HOURS LS digit decimal counter module (HP Assembly 5212(x)-4A) to reset to a count of '1' instead of '0', simply modify that module so that R23 connects to the base of Q2 instead of to the base of Q1.

And finally, for item 4:

The "Transfer Line" signal to the counter modules controls the modules "storage" function (necessary for a counter to maintain a stable display while the modules are counting). But there's no reason to use a storage function when operating as a clock -- we can simply view the count while it's incrementing, and it makes the modification simpler

To disable the "Transfer Line", I cut the wire from the driver that drove pin 5 of the Decimal Counter card-edge connectors. This line will then float at about-12V, and the storage feature of the Decimal Counter modules will be disabled).

So...that's essentially it!

[I did add various switches and buttons to give me some functions that I wanted (such as turning off the display, but continue running the clock, to save on "wear and tear" of the nixies). But the modifications listed above are the main mods to the basic counter function.]

Here's the top view of the modified counter...

...with the modified decimal-counter modules:

And a view under the chassis. The HOURS reset circuit is wired between two of the edge connectors towards the right.

The counter, before I made an overlay for the front panel...

...and after:

(Martini time!)

From left to right, front panel controls are:

Toggle Switch: Mode, RUN/SET-Time
Push Button: Increment clock when in SET-Time mode
Toggle Switch: FAST/SLOW increment speed when in SET-Time mode
Rotary Switch, 3 Position. Selects Display Format: H:M:S, H:M, Display OFF (but clock still counting)
Push Button, Reset SECONDS count
Pot with Power-On switch. Only the power-on switch is used.

Other Notes:

1. HP Manuals are great. I've always found their "Principles of Operation" section to be an excellent description of how their equipment operates -- very useful and well worth checking out.

2. To use a 5212(x)-4A module in slot A19 of the 5233L counter, clip wire going to pin 8 of that slot's card-edge connector. (I no longer recall which module was originally in this slot).

3. You can use the "decimal point" neon lamps in the modules as separators to differentiate between hours, minutes, and seconds digits (refer to photo of my counter).

[Note: the 5212L-4A modules in the 5233L counter do not have neon lamp decimal points within the modules themselves (unlike the 5212A-4A module). Instead, the neon lamps are mounted on a separate small PCB that runs underneath the modules, next to the front panel.]

4. You could probably make this a 24-hour clock as follows:

Do not modify the Hours LS digit to reset to 1.
Change the Hours Reset Circuit to use the following 2 bits instead of the 3 bits shown in my schematic above:

Hours LS Digit pin 13
Hours MS Digit pin 9

(I haven't tried this, so take this recommendation with a grain of salt.)

5. Pin 7 is the Clock input pin for a 5212(x)-4A module. It is to this pin of the first module (the least-significant digit of the "seconds" modules) that you connect the 1 Hz time reference (and the faster clocks, when in SET-Time mode).

6. In an HP counter you should find a number of identical divide-by-10 modules that are used to divide down the time-base reference frequency (HP p/n 5212A-65C). It is on the card-edge connectors of these modules that you can find the 1Hz, 100Hz, and 1KHz signals (as well as others, should you need them).

Note 1: These boards have pin 7 as their input and pin 5 as their divide-by-10 outputs.

Note 2: Although my block-diagram above has 1 KHz as the "fast" set-time frequency, I might have actually 10 KHz.

7. After I'd mucked around with making an overlay and installing it on the front panel, the counter started "double-counting" (it would increment twice in one second -- on the rising and the falling edges of the 1 Hz clock). It turns out that CR9 on the first nixie tube module had opened up. HP's p/n for this part is 1910-0015, but unfortunately it didn't list the manufacturer's part number. I replaced it with a 1N4148 -- and although this new part is silicon, rather than germanium of the original diode, it fixed the problem.

8. A note regarding my "Reset Seconds" pushbutton. This button simply zeroes out the MS and LS seconds digit (and it also keeps the other digits from counting). But if one does this when the seconds count is greater than '19', the minutes count will increment by one. This isn't a big deal for me, and actually, it's kind of useful when setting time. Here's what I do:

I'll run up the time to the current time (per WWV), making sure that there are at least 19 seconds on the clock.
Then I'll flip the RUN switch to RUN and depress and hold the Reset Seconds button. This will set the clock to the next minute and zero the seconds.
I then wait, with button depressed, until WWV hits the minute mark before releasing the button.
The clock is now within a fraction of a second of WWV!

9. The +20VDC rail within my counter was way off (it read +31 VDC). Some probing revealed that a couple of transistors, as well as a zener diode, in the voltage regulator circuit had blown. The two transistors were both germanium PNP parts (one was an HP 1850-0062, which crosses to a 2N404A, and the other was an 1850-0105, for which I cannot find a cross reference). I replaced these both with 2N3905 transistors (PNP silicon). I replaced the blown zener with a 6.8 volt one, and, when finished, the voltage read +19.98 VDC. The voltages across the 2N3905 transistors are well within their range, and they aren't getting warm, so I believe everything should be copacetic.

Final note...this posting was written not to give detailed instructions for modifying an HP counter to be a clock, but rather to generate ideas and inspiration. If you have an old HP counter kicking around somewhere, consider giving it a try!

- Jeff, K6JCA

Improving AM Performance of the Heathkit MT-1 Cheyenne Transmitter

2009-12-10T14:19:00.000-08:00

[Update (2 January 2010): New information on my Cheyenne can be found here]

I picked up this transmitter at a swapmeet earlier this year. Although the front panel was in nice condition, the topside of the chassis itself had oxidized quite a bit (as had the cabinet, which was in dire need of rust removal and repainting), and it was not very appealing. Never the less, the price was right, and I thought it might be a fun project to get on the air during the cold winter months!

The Cheyenne -- case removed for repainting and
needing an original knob (hint hint) for the Drive control.
(Click on image to enlarge.)

Hmmm...under the hood, not so pretty

The Heathkit Cheyenne transmitter was a mobile AM and CW transmitter that Heathkit marketed in the late 50's and, I believe, early 60's (its matching receiver was the Heathkit Commanche (MR-1)). With a design including a single 6146 PA, 12AX7 Mic amp, and 6DE7 as a "controlled carrier" modulator, it is similar (although not identical) to the later DX-60 series transmitters.

I powered up the transmitter with an HP-20 power supply and quickly discovered that its audio in AM mode left quite a bit to be desired -- noticeable distortion and a restricted audio passband. So I wondered...what could I do to improve its performance?

Well, the first thing to do: check to see if someone else has already been down this path. Unfortunately, a google search revealed no internet articles for improving the Cheyenne. But, because of the similarities between the Cheyenne and the DX-60, I wondered if I could apply any of the DX-60 modification articles to the Cheyenne...

First Steps...Make the Cheyenne more like a DX-60...

Electric Radio magazine has several very interesting articles by Bill Breshears, WC3K, on improving the DX-60 audio. I thought they might be a good starting point for modifying my Cheyenne, but to do so, I'd first need to correct those few differences between the Cheyenne's audio/modulator stages and the DX-60's stages.

Upon comparing the Cheyenne schematics with those for the DX-60B, the significant differences in the modulator section seemed to be:

The DX-60B's 6DE7 Cathode Follower has a 33K ohm resistor from its cathode (pin 9) to ground. This resistor was lacking in the Cheyenne (and indeed, its lack prevents the AC signal on the Cheyenne's cathode-follower cathode from going below about 50 VDC).
The grid of the first stage of the DX-60B's 6DE7 modulator (pin 7) has a 22 Meg ohm resistor to ground, compared to the Cheyenne's 10 Meg, and this grid is driven by the previous 12AX7 stage via a 5 nF cap, instead of a 510 pF cap in the Cheyenne.
The DX-60B's PA screen voltage is driven by the 6DE7 Cathode Follower through a 47K ohm resistor paralled with a 0.1 uF cap. The Cheyenne uses a 10K ohm resistor and a 0.25 uF cap.

I incorporated the changes in first two items above (although I used a 6.8 nF cap in lieu of 5 nF in step two, because that's what I had in the junk box). I left the third item for later, until I could incorporate the Carrier-Level adjustment pot that WC3K described in his articles.

One of my goals was to drive my AL-811 linear amplifier with the Cheyenne. I don't feel comfortable running this amplifier in AM mode at more than about 100 to 120 watts carrier output power. For this level of power output from the linear, I needed the Cheyenne's idle-carrier power output to be to be in the range of about 9 watts or so.

I initially incorporated the 47K ohm resistor in step 3 above (keeping the cap at 0.25 uF) and added a 25K ohm pot in series with it to the PA Screen Grid (the 0.25uF cap paralleling both) -- similar to the carrier control pot described by by WC3K in his DX-60 mods. Unfortunately, I felt I had a bit too much drop in power, and I instead replaced the 47K with the original 10K. (This would later change again. See below...)

The new pot (mounted conveniently in the rear-panel's Key Jack hole (after all, who needs a key for AM operation?) allowed me to easily adjust carrier level. But during testing I wasn't satisfied with audio performance -- there was a still a bit of "fuzziness" on the audio (pointing to distortion) that bugged me.

Examining the audio chain, it became quickly apparent that part of the problem was with the 6DE7 "controlled-carrier" modulator itself. This modulator adjusts the carrier level such that the carrier level is low for low-level signals and higher for high-level signals. To accomplish this "dynamic" carrier-level adjustment, the first stage of the 6DE7 modulator, in addition to being an AC amplifier, "clamps" the input AC signal on its positive peaks, thus causing an additional DC voltage to be impressed across the coupling cap (that couples the signal from the second 12AX7 stage to the input grid of the modulator). This DC voltage is proportional to signal level and thus drives the modulator grid (6DE7 pin 7) more negative with higher audio levels.

As this grid is biased more negative (relative to the grounded cathode), the tube conducts less, reducing the DC plate current. The plate voltage goes up, thus raising the grid voltage on the next stage (cathode-follower) and consequently, of course, its cathode voltage.

As this cathode goes up, the PA Screen Grid voltage goes up, and more carrier appears at the output.

BUT -- the key here, and the source of the distortion, is the clamping action at the grid of the first 6DE7 section. This clamping action essentially flattens the positive peaks of the audio signal at this grid, which in turn results in flattening of the "troughs" of the modulation on the output RF signal.

This flattening of the audio signal is easily observable at the 6DE7 with a scope (monitor the plate of the first stage, for example), and is quite obvious on a 1 KHz test signal. Not good. My rule of thumb is...if you can see the distortion, you can hear it.

I also had a problem in which, as I tried to adjust the Cheyenne's audio level towards 100 percent modulation, I'd get compression (i.e. distortion) on modulated RF envelope "peaks". Again, this was readily apparent by comparing the modulation on the RF signal with the audio signal driving the modulator.

Here's photo showing both of these two distortion mechanisms (exaggerated to make it clearer) . The top trace is the modulator output. You can see the clipping on the largest negative peak due to clamping by the modulator input grid. The bottom trace shows peak compression on the output RF envelope, which you can see by comparing the different levels of the positive peaks of the modulator output with the peaks of the RF envelope -- they're all the same level!

The Next Step...Improving the Cheyenne's Audio...

The fuzziness on the audio was just enough to make me want to keep working on the transmitter. One source of this distortion, as discussed above, was from the clamping action of the "controlled carrier" modulator in order to dynamically adjust carrier level.

Hmmm...suppose I eliminated this clamping action and thus the distortion that it created? Would audio be improved?

Why not? I only needed a carrier to be somewhere in the range of 8 to 12 watts to drive my linear. There's no reason why the 6146 PA in the Cheyenne cannot handle this. In other words, why not remove the "controlled-carrier" feature of the modulator (and thus the distortion that it introduces) and keep the carrier at a fixed level?

And I wanted the carrier level to be adjustable so that I could adjust the level to give me my 100 watt "sweet-spot" output from my linear.

One way to get around the "control carrier" feature is to bias the first stage of the 6DE7 modulator so that there is a fixed negative grid-to-cathode voltage that is large enough to prevent clipping on the positive peaks of the incoming audio, yet provide sufficient carrier to drive the linear. I already had a 25K pot mounted on the back panel of the chassis that I had intended to use to adjust carrier level (and had been wired in series with the original 10K power resistor to the PA screen grid -- see discussion above). I wired it instead into the cathode of the first stage of the 6DE7 so that I could adjust its operating point.

Through a process of iteration, I adjusted the PA screen grid resistor value and the position of the 25K pot so that, for the carrier output power that I wanted (8-10 watts), the cathode of the first 6DE7 was at a high enough voltage that full-modulation audio wouldn't be clamped by the grid. Thus, the PA screen grid resistor (from V6 pin 9) was changed from the original 10K ohms to 50K ohms. Although I used a robust power resistor (it was in the junk box), there's no reason why, say, a 2-watt resistor couldn't be used. And you can play around with the value of this resistor -- lower values of resistance will increase the maximum carrier power, while higher values will lower the maximum carrier power (maximum carrier power occurs when the 25K pot is set to its maximum resistance).

[Important Note: there's a trade-off when selecting the value of the PA screen resistor: for a given carrier output power, lowering the PA screen grid resistor value means that the resistance of the 25K pot in the cathode of the first 6DE7 section must also be lowered to maintain the same carrier output level. This in turn will bring the cathode voltage closer to the grid voltage (which is essentially at 0 volts), which means that it's more likely there will be audio distortion introduced at this 6DE7 grid due to grid "clamping" the positive peaks of the audio signal. I found that a 50K ohm PA screen grid resistor worked well for my application.]

I found that the value of the pot is about 8K-9K ohms for about 9 watts carrier (no modulation) RF output from the Cheyenne. Given this value of resistance, I added a 1 uF cap in parallel across the pot to bypass it for audio frequencies. (Note: This cap can be made larger, if it's desirable to run the Cheyenne at lower power (and thus a lower potentiometer resistance, which means you need a larger cap to maintain the low-frequency cutoff), but increasing its value will also increase the amount of time that it takes for this stage to reach its bias point each time PTT is pressed.)

By the way, with these mods made and the Cheyenne set for about 9 watts carrier output (no modulation), I measure the following DC voltages during Transmit:

V6.5: 6 volts (6DE7, first cathode)
V6.2: 110 volts (6DE7, second grid)
V6.9: 200 volts (6DE7, second cathode)
V4.3: 52 volts (6146, screen grid)

These changes gave me the ability to control the output carrier power from about 4 watts to about 12 watts. Note -- at low powers there may still be some peak flattening at the grid to the first 6DE7 stage (this occurs when the cathode-grid bias voltage is less than the audio peak voltage at the grid), but I've found that there's no limiting when the 25K pot is set to give me 8-12 watt carrier power output.

OK! Now that I had the distortion reduced, I next tackled the frequency response, which was a bit too restricted in the stock Cheyenne.

This was accomplished (in addition to the changes above) simply by :

Changing the 0.001 uF cap feeding the grid of the first 12AX7 stage to 0.01 uF.
Changing the 510 pF cap feeding the Audio Level pot (from the plate of the first 12AX7 stage) to 0.01 uF.

This gave me an audio passband with -3dB break-points at 100 Hz and 5 KHz.

The new schematic:

(Modulator Modifications -- Click on schematic to enlarge...)

Other problems:

1. 6.3 VAC reading low on DVM at the terminal strip: only about 5.5 VAC (causing the relay to chatter):

Bypassed the fuse in the filament line with a wire soldered to the fuse-holder's terminals (this fuse is not needed, after all, the power supply is fused, and this fuse added a few additional tenths of a volt of voltage drop).
Cleaned the Function switch contacts.

2. The 0.02 uF cap attached to V6 pin 1 (600v bypass) "popped." Replaced.

3. AC Hum on AM signal which gets louder as mic gain is increased. The PTT signal of the Cheyenne's mic jack directly keys the Cheyenne's relay, which is powered by 6.3V AC. A mic with wired to a 4-pin plug to mate with the Heathkit mic jack shouldn't have an issue with this, because, assuming the mic and its cable have separate grounds for the PTT return and the audio return.

Unfortunately, a number of my mics have common PTT and audio grounds (and are terminated with PJ-068 plugs), which means that the 6.3VAC on the PTT line runs on the same ground line as the audio return and thus contaminates the audio signal with AC hum.

I really didn't want to rewire a mic with a Heathkit-compatible plug -- I preferred to keep them terminated with PJ-068 plugs so that they're interchangeable among a number of my transmitters. Instead, I made an adapter with a PJ-068 compatible jack and a 4-pin Cheyenne compatible plug. Because the common ground would create a hum problem, I decided to have the PTT switch control a DC, not AC, signal, thus removing AC crosstalk from the common return line.

To do this I added a second, 5VDC relay, and rectified the 6.3VAC filament voltage to provide the voltage to drive this relay (see the schematic above). The mic's PTT button now switches this DC voltage. The new relay then switches the 6.3VAC signal to the original Cheyenne relay.

The new relay also has an additional benefit -- I wanted some way to mute an external receiver during transmit as well as key an external amplifier, and the extra contacts on the relay now provides these functions. (A previous owner of the Cheyenne had rewired the 6-pin connector that connects to a receiver (such as the Commanche) and several of these pins were left unused, so I brought these two new signals (Receiver Mute and Amplifier Key) to these unused pins on this connector).

(Additional note: the 6.3 VAC relay has a coil resistance of only about 8 ohms. This means that it draws about 0.8 A when ON. The only 5VDC relay I could find in my junk box has contact ratings of 1A at 30 VDC. OK, 1A is greater than 0.8A, but personally, I'd prefer a bit more margin. It seems to be working well so far, though.)

(Mounting of the 5V relay)

4. The SPOT switch did not work in STBY mode. Incorrectly rewired by someone in the past, I connected it to pin 5 of the relay (300V when not transmitting, although it really should go to pin 4 of the relay (per the schematic), but the remaining wire wasn't long enough, and pin 5 is a good compromise).

5. VFO tracking way off. Re-adjusted, but during this readjustment I discovered that the bottom-end of 80 meters would quickly diverge despite the rest of the band tracking well. Because I intend to use the transmitter for AM only, I decided to leave well-enough alone and I adjusted the bandspread to track the VFO dial over the range of 3.7 - 4 MHz.

Still to be Resolved:

1. Oscillator: the 1.8 MHz fundamental at the plate of the oscillator 6AU6 (that's later doubled to provide the 80 meter signal) looks terrible, as can be seen in the top trace below (the bottom trace is the RF output (sampled via an attenuator)):

I'm surmising that the signal on the oscillator plate looks this way because the 8.5 uH inductor in the 6AU6 plate circuit is differentiating a 1.8 MHz plate-current pulse-train from the 6AU6 (after all, v = Ldi/dt). But is this the way it really ought to look? I've no idea.

Never the less, the transmitter seems to perform OK and I cannot find anything obviously wrong in the oscillator circuit, so I'm going to reserve judgment...

(Additional note: The waveform at the grid of the oscillator looks great -- a nice sine wave. Just the plate signal looks weird. Ought to have a resonant circuit to make it look nice, I think.)

2. There is a bit of audio roll-off from about 1.5 KHz (down 1 dB from 1 KHz) to 5 KHz (down 3 dB from 1 KHz). The first place this roll-off appears is at the plate of the second 12AX7 stage (yet it looks fine at this stage's input grid). I've yet to identify the cause -- I suspect it might be roll-off caused by the RC network formed by the AUDIO pot and the 12AX7 grid capacitance at pin 2, but...

3. I'm not sure if this is a problem or not, but carrier power does increase a bit as I approach full modulation, even though I've disabled the "control-carrier" feature of the modulator. I'm not sure what the cause is. Perhaps a non-linear PA Screen Grid transfer function? Or...?

4. VFO drifts.

Tuning up the Transmitter:

It's important that the transmitter's loading be properly adjusted. If the loading is too light you'll get peak compression on the RF envelope. I find that adjusting loading for peak power in CW mode actually puts it about where it needs to be for AM.

Here's the procedure I use. It seems to work.

Rotate LOAD and AUDIO controls fully counter-clockwise.
Place function switch in GRID position. Press PTT and adjust DRIVE for 3 mA (or for peak reading if 3 mA cannot be reached).
Place the function switch in PHONE and the meter switch in the PLATE position. Press PTT and dip the plate using the FINAL control.
Switch the meter switch back to GRID. Press PTT and ensure grid drive isn't exceeding 3 mA. (Important note: I've found that as I rotate the DRIVE control through 360 degrees, I hit the 3 mA level at four positions. And for two of these four locations, the output power is greater than for the other two locations, despite the equivalent 3 mA drive level. When performing the final DRIVE adjustment, be sure to select one of the two DRIVE positions that results in greatest output power (at 3 mA drive)).
Switch the function switch to CW. Press PTT and advance LOAD to peak the power output. Dip plate current again, just to be sure.
Switch the function switch back to PHONE. Press PTT and adjust the new CARRIER LEVEL pot (on the back panel of my Cheyenne) to the desired carrier power out (no modulation). Then, while talking into the microphone and monitoring the RF envelope on a scope, advance the AUDIO control until the "troughs" of the modulation envelope are just on the cusp of flat-lining. If you find that the peaks of the envelope reach their max level before the troughs reach their min level, you have too much carrier and you should back down the CARRIER LEVEL pot -- you're just wasting power.

You're done!

Other Notes:

1. One goal was to make my modifications without drilling new holes in the transmitter, so that if someone, at a later date, wished to return the transmitter to its original condition, they could. Fortunately, it was fairly easy to add additional terminal strips using existing screws, and the key-jack hole on the back of the chassis was an ideal place to mount the Carrier-Level pot.

Additional terminal strip for new components. No holes drilled!

2. For proper AM operation, the Loading control must be adjusted for peak power out (and perhaps even a bit beyond, to be on the safe side), otherwise the positive modulation peaks will be greatly compressed and your audio will sound lousy.

3. The 1 uF cap across the new carrier-level pot can be made larger if it's desirable to run the Cheyenne at lower power (and thus a lower potentiometer resistance) and to keep the low-frequency cut-off. But increasing its value will also increase the amount of time that it takes for this stage to reach its bias point when PTT is pressed.

4. At higher carrier levels the Cheyenne has somewhat less measurable distortion at close-to 100% modulation than it does at lower carrier levels (as long as you aren't exceeding the capabilities of the PA on voice peaks, of course). I attribute this to non-linearities in the PA's screen grid transfer function (hypothesized, but not proven).

5. Here's a transfer curve that I've made, using my Cheyenne transmitter, showing RF output voltage (attenuated by my RF "sampler") versus PA Screen Voltage (note: I'm using a 6293 tube in lieu of a 6146). Test conditions: PHONE mode, 80 meters, no modulation.

You can see the curve bending at the high voltages (this results in compression of RF envelope peaks). It looks pretty linear at lower voltages (the slight burbles are most likely due to measurement error (of either screen voltage or RF amplitude) on my part).

(Click on image to enlarge)

6. WC3K's articles in Electric Radio (regarding the DX-60, see below) describe a neat modulation monitor using an LED. There's no reason why this can't also be used with the Cheyenne. I didn't install it, because I didn't want to drill a hole in the front panel and, besides, I use a scope to monitor my modulation. But I recommend taking a look at it. (You can find similar circuits in some of the web sites discussing DX-60 mods, too.)

7. A reminder: Update (2 January 2010): New information on my Cheyenne can be found here.

Resources:

Heathkit MT-1 "Cheyenne"Information HERE

Heathkit Schematics HERE

Unfortunately, I couldn't find any information regarding modifying the Cheyenne on the web. However, its design is similar to the DX-60, and there are articles that discuss improving AM performance of the DX-60...

Electric Radio articles on DX-60 improvements:

"Fun with a DX-60," Bill Breshears, WC3K, Electric Radio, Issue 133, May, 2000
"More Fun with a DX-60," Bill Breshears, WC3K, Electric Radio, Issue 138, November, 2000

Websites with DX-60 improvements or discussions:

WC3K Improvements (note: the 50 ohm pot should be 50K ohms)
K4TAX Improvements
KS3K Improvements
N4JK Modifications
K4TLJ Modifications
LA8AK

Let's see...where did I put that screwdriver?

Standard Caveat...

I hope you find this information useful, but please, use these modifications at your own risk -- although they worked for me, I cannot guarantee that they'll work for you. (After all, I could have made a mistake in transposing them from my lab notebook to this post.)

If you do find any errors, or if you have any questions, please let me know. Thanks!

- Jeff, K6JCA

R-105A/ARR-15 Receiver

2009-10-29T09:05:00.000-07:00

[Update (4 January 2010): Additional info on modifying the R-105A to improve selectivity can be found in my new blog posting here.]

I picked up this receiver, along with a companion ART-13 transmitter, a couple of years ago. Both are in "well-used" (beat-up) condition, but...what the heck. I'd been looking for an ART-13, and the ARR-15 intrigued me. And no, the tuning knob isn't original.

Here's a picture of them in the radio operating position of a military plane. (Photo is from this website: 51H-3.)

(Click on image to enlarge.)

Although I'm well familiar with the ART-13 transmitter (having disassembled one for parts back when I was in high-school), I've never seen (nor heard of) the R-105A receiver. It's the military version of Collins 51H-3 receiver, manufactured post-World War II (mine has a 1951 contract date). And, apparently, it was Collins first remotely-tunable receiver (tunable to 10 preset frequencies).

Although intended to be used on 10 preset frequencies, the receiver can also be tuned the "normal" way via a tuning-knob and band-switch on the front panel. Frequency coverage is 1.5 - 18 MHz in 6 bands, and modes are MCW (AM) and CW.

The R-105A is designed to be powered from 26.5 volts DC, and it uses an internal dynamotor (DY-34) to convert this voltage to 220 VDC for tube B+ voltage. If using the dynamotor, I believe an external power supply should be rated at 15 amps, 26.5 VDC.

My radio did not have the dynamotor installed. Instead, a previous owner had wired the B+ line to one of the spare pins on the back connector. My receiver's power requirements are:

26.5 VDC (filaments/motor): 1.4A normally, 5A (or a bit more) when Autotuning.
220VDC (B+): about 70 mA.

Here are some photos. Despite the relative shabbiness of the exterior, the interior is actually in nice shape.

(R105A, Top View)

(R105A, Bottom View)

(R105A, Right Side View)

(R105A, Left Side View)

Getting It Up and Running...

OK, the only documentation I had was a schematic that I downloaded from the web (see "Resource" section, below). The radio had no dynamotor, but the previous owner had brought B+ out to pin 18 of the rear connector. So I attached a 220 volt supply between pins 18 and 9 of the rear connector ("plus" to pin 18), and 26.5 volts between pins 17 and 9 ("plus" to pin 17). I attached a pair of headphones and an antenna, then switched on the power supplies, turned on the radio's front-panel switch, and...

Nothing. The dial-lights were lit, but I couldn't hear anything -- it was as if the receiver was dead.

I looked at the schematic again and noticed that the resistors in the cathodes of the RF Amplifier and the First IF Amplifier weren't grounded, but were instead going to pin 3 of the rear connector. Clearly they needed to be connected to something (such as ground), but what exactly should this be?

One of the websites I visited mentioned that, in CW mode, the front-panel Gain pot is used to control RF, rather than AF, gain. Hmmm...RF gain as in, perhaps, the cathode of the RF amplifier? Ah ha! A clue!

I noticed in the schematic that there was one section of the gain pot, R139C, that, when the radio was in CW mode, was connected to pin 20 of the rear connector. Could it be as simple as connecting pin 3 to pin 20 on the rear connector?

Yes! I connected these two pins together, applied power, and...signals!!!

There were still some issues, though. I could hear distortion on AM signals, and I could see that, for whatever reason, there was way too much gain -- so much so that the AF Amplifier was being driven into distortion for reasonable-level signals.

When I looked at the R-105A schematic that I had downloaded from the BAMA site, I quickly realized it did not match my receiver. In fact, that schematic is for the R-105 (non-A) version, and there are some significant differences, particularly in the Audio stages. So I traced out my receiver's circuit from the detector up to (but not including) the AF Amplifier. Here it is:

(Click on image to enlarge)

Regarding the distortion and gain issues, my primary suspects were the limiter and the AVC circuits. But I looked at my schematic and quickly realized there were some strange things in the design and that I had no idea how the limiter and AVC were really supposed to function. I poked around with a scope and DVM for a few days but didn't make any headway. What I needed was a good description of how these circuits were supposed to operate. Usually the military tech manuals contain some sort of theory-of-operation descriptions...it was time to try to round one up...

After a bit of searching, I found someone on the web that could sell me a manual reprint (see "Resources" below), and I ordered it. It proved to be quite useful...

The first thing that I discovered upon reading it is that pin 3 of the Limiter stage (V110), during normal operation, should be higher in voltage than pin 8 of the same tube (that is, both diodes are conducting). In my radio pin 3 was lower than pin 8 (despite the fact that the plate voltage of V105A was higher than V107A) and the diode of V110B wasn't conducting all of the time. Oh oh. Cap C133 looked fine -- must be a leaky 12H6. Unfortunately, I didn't have a spare tube in my tube-stash, so I made a solid-state replacement using an octal plug, two 1N4006 diodes, and an 80 ohm, 3 watt resistor (to mimic the tube's filament load -- I made this using 3 power resistors I found in my junkbox). The octal-plug was wired as follows:

80 ohm resistor between pin 2 and 7
1N4006 Anode to pin 3, Cathode to pin 4
1N4006 Anode to pin 5, Cathode to pin 8

I plugged it in and...the voltages were now OK! (This mod should suffice until I can find another working 12H6 tube.)

But there was still a gain issue -- during modulation peaks, loud signals would flat-top at the output of the AF Amplifier. (Note: the front-panel gain control does not control the level of the signal fed to the AF amplifier, it actually controls the gain (via attenuation) right at the headphones. Thus it's reasonable to expect the AF amplifier to operate at a high level (to get the best dynamic range), but...it should never go into clipping!)

I spent quite a bit of time exploring the AVC and audio stages...was there too much gain in the audio? Was there not enough gain (or leakage) in the AVC circuit? Or...?

Although there's quite a bit of gain in the audio stages, it looked to me, from the component values and from what I was measuring, that the gain I was seeing was reasonable (and I reduced the gain of the AF driver as much as I could by setting R156, an internal pot, to its max value). I checked the AVC line for leakage or loss (the AVC line drives the grids of the RF amplifier the 1st IF Amplifier) -- it looked fine. Finally, after much poking around, the only explanation I could come up with was that there simply wasn't enough AVC control-voltage being developed to keep loud, highly modulated signals from clipping.

How could I develop more negative AVC voltage?

Looking at the schematic for the AVC circuit, it is is unlike any I'd seen before. Although there is a diode detector (V106A), this is only used to change the signal-level threshold at which the AVC begins operating, rather than, as is typical, developing the AVC voltage itself.

Instead, it is the second section of V106 (V106B) that actually develops the AVC voltage.

It does this by acting as a variable load on the AC-coupled IF signal (coupled to the tube via C123). If there is no AGC action, this IF signal sees R121 (1 Meg) as its load, and R125/C129C low-pass filter the signal across this load.

With small signals, the cathode of V106B sits at about 17 volts (this level is set by the voltage divider formed by R132, R122, and R133). For signals whose amplitude, at the plate of V106B, is less than 17 volts, the tube is in cutoff and, effectively, out-of-circuit. Thus the IF signal only sees R121 as its load, and because the IF signal is AC-coupled and R121 is unchanging, the AVC voltage, after the IF signal has been low-pass filtered, is 0 volts. (That is, the low-pass filter is essentially an "averager", and the average of an AC signal that is symmetric and centered on 0 volts is...0 volts.)

If the signal amplitude on the plate of V106B exceeds the voltage of the cathode, V106B begins to conduct (the amount of conduction is determined in part by the cathode-grid voltage: note that the grid is tied to ground). When the tube conducts, it acts like a finite-valued resistor in parallel with R121, the 1 Meg load resistance, and thus the load resistance seen by the IF signal (coupled via C123) is lowered. Because the tube only conducts on postive peaks, the IF signal sees this smaller load (and thus more attenuation) only during its positive peaks, but not during the remaining part of this signal's cycle. Thus, there is more attenuation for positive peaks than for negative peaks.

Because the positive peaks are attenuated compared to the negative peaks, the "average" of the signal is no longer 0 volts, but instead it is a negative voltage. And this is the AVC voltage.

V106A is used to lower the cathode voltage for strong signals to drive the AVC voltage more negative -- if the cathode is lower than 17 volts, the tube will begin conducting at a lower positive signal amplitude, and thus more of the positive peaks of the IF signal will be attenuated compared to the negative peaks, and thus the AVC will become more negative.

Essentially, V106A acts as a diode detector, detecting the IF signal coupled to it via C132 and developing a negative voltage which, when fed to the cathode of the second stage of V106 (via R123), subtracts from the 17 volts that is normally there (fed to the cathode of V106B via R126). C186 filters out the high-frequency IF signal, leaving only its negative audio envelope.

I needed to develop more negative AVC voltage during loud signals. After experimenting, I was able to get reasonable performance with this simple mod (which can be easily backed-out if one is a purist and wishes to keep their receiver in original condition) :

Parallel R123 (470K) with a 47K resistor.
Parallel C186 (470 pF) with a 4.7 nF capacitor.

This modification drives the cathode of V106B lower (on average) than occurs with the stock 470K resistor in R123 (because R123 is smaller, there is less voltage "lost" across it, due to the voltage-divider action that takes place with R126, and hence the cathode of V106B is driven lower (but never less than about 0 volts).

The change in the value of C186 matches the change in R123 and keeps unchanged the time constant of the filter formed by R123 and C186 (which filters out the IF frequency, leaving only the modulation envelope).

It seems to work well. In my listening tests (and measuring with a scope) there is certainly less distortion with the mod than without it.

That's it! Besides that, I haven't changed anything else in the receiver.

Other notes:

Althought the receiver really isn't designed for SSB use, it can be used in that mode, although tuning is a bit too fast.

The autotune is really very cool! (There are 10 channels you can preset.)

The IF is quite broad. It reminds me of using a Command Set receiver.

To Mute the receiver during transmit, add an SPST switch between pin 3 and pin 20 on the rear connector. This switch should be closed during receive and open during transmit.

With its octal-tube sockets and well laid-out design, the R-105A is a real pleasure to work on, especially when compared to typical ham boatanchors in which components are often buried under other components, making access difficult, if not impossible.

Resources:

Schematics here (BAMA site). Yes, they are small and difficult to read. But...they're the only schematics I could find on-line, and they're better than nothing at all. IMPORTANT NOTE: Although the BAMA site lists these as being schematics for the R-105A, they are actually for the earlier R-105 (non-A) version! There's a crystal rectifier detector shown in the schematic in lieu of a detector implemented with 1/2 of V105 (as my R-105A has). And V107 is shown as a 12SJ7 instead of a 12SL7. (By the way -- there's a mistake, too, in the schematics: they incorrectly show R107 connected to the same line as R111 (First Mixer's Cathode resistor). Instead, R107 should connect to B+. And I've no doubt there are other differences...)

AN/ARR-15A feature summary.

51H-3 Good information and a great picture of an ARR-15 / ART-13 pair aboard a P2V anti-submarine patrol bomber.

More pictures here.

Tech Manual: AN 16-30ARR15-3. [You can purchase reprints of this manual (as of 30 Oct 09) from WA5CAB.]

Rear-Connector Pin Assignments (traced from the schematic: click on image to enlarge):
And finally, a reminder that my later post (on improving the R-105's selectivity) can be found here.

Standard Caveat -- take everything I've written with a grain of salt. I could have easily made a mistake.

Thanks!

- Jeff, K6JCA

KW Atlanta Transceiver

2009-10-18T09:13:00.001-07:00

This cute little radio followed me home from last weekend's De Anza Swapmeet. It's an "Atlanta" transceiver manufactured by KW Electronics, Ltd., of Dartford, England.

KW Electronics manufactured radio equipment for the British market. I believe the Atlanta transceiver was their one foray into the American ham marketplace and was sold here back in the early 70's (although I must admit that I never heard of the company at that time).

As you can see from the photo, the radio isn't in "original" condition. There's a toggle-switch just to the right of the band-switch (discussed below). There's also an additional DC "accessory" connector in the upper left-hand corner of the Power Supply front panel, and I'm not sure if the meter is original, or not (I suspect it's not).

The radio uses two 6LQ6 sweep tubes in its PA (also compatible: 6JE6 tubes). PA final voltage is spec'd at 800 volts, but mine measures 700 volts in receive, and 650 when loaded in transmit mode (I don't know if the power-transformer is original, or if it has been replaced).

So, given the lower PA voltage, the output power isn't quite as high as I would have expected it to be, but it suffices.

On the receive side there can be "popping" on the starting edge of loud signals. The AGC is audio-derived (rather than being derived prior to the detector) and it has a rather slow attack time, so some amount of leading-edge popping is to be expected.

But, despite these small problems, overall it's a nice package.

Here are some of the issues I ran into getting it on the air...

1. No Power Switch -- Wired Permanently ON. There was a "goof plug" on the front panel, just to the right of the band-switch. I opened up the radio and discovered that, at one time, there had been a power switch mounted to the back of the AF Gain potentiometer, but it had been removed by someone. Apparently they then drilled a hole in the front panel for a toggle switch, but by the time the radio got into my hands the toggle switch had been removed and the radio wired to be permanently On. I removed the "goof plug" and put a toggle switch into the existing hole.

2. Very low speaker volume -- someone had added a cap in series with the speaker and also ran a separate ground for the speaker from the radio to the power-supply unit (which they left unconnected). Why, I don't know, but I suspect they made these two mods to reduce speaker hum. But the cap was much too small, only 1 uF or so (thus presenting a very large series-impedance at AF frequencies). I removed it and wired the seperate ground to the speaker, and it works fine now.

3. Intermittent receive signal strength. Receive signals would sometimes be loud, and other times weak. If I touched the rear panel, I could make them fluctuate: clearly an indication of a bad connection somewhere. After hunting around, I finally found a lose connection at the bottom of L2 (inductor in the ouput pi-network). A very awkward location. Luckily, I had a very narrow soldering iron, so I could get to the bottom of the coil and repair it!

4. Low Power Output -- only 50 watts or so. I brought the power up by peaking transformer T1 per, I thought, the instructions in the manual (they're a bit ambiguous). However, carrier suppression was now terrible. I was able to get better carrier suppression, but I had to change T1's alignment procedure. Here's what I did:

T1 has two cores. Move the core closest to the chassis all the way to the chassis-side of T1. Let's call this the "bottom" core.
Transmit and insert some carrier, then adjust the top core for max TX signal.
Stop transmitting, and then adjust the bottom core for max RX signal.
Repeat, if necessary.

5. Poor Carrier Suppression. First, I adjusted the frequencies of the two carrier-oscillator crystals so that the bandwidth in both modes (when operation, say, on 80 meters) was close to identical (BW about 300 - 3100 Hz, measured using a white-noise source fed into the mic inputand a spectrum analyzer on the output). This put the carrier level for both sidebands at about the same level (when viewed on a spectrum analyzer). Then, with the Carrier Balance knob pointing straight up, I adjusted C114 for minimum carrier in both sideband modes. Then use the front-panel pot for fine-adjusting. You might need to do this several times before getting a good null. (Note, I'd tried to null carrier by nulling with the pot first and then adjusting C114. I could never get good suppression this way.)

Ongoing Problems...

1. Carrier does not remain suppressed. Don't yet know why...

(Update, 25 October 09: While poking around, I discovered that the two 100K resistors in the balanced modulator (7360) bias-network circuit had drifted an enormous amount -- one measured 232K, the other measured 335K! I ran a quick calculation (using the measurements of voltages that I'd made) and discovered that, if they had been 100K resistors, they each would have been dissipating more than 0.5 watts. Not good (given that they're 0.5 watt resistors), and perhaps the cause of the enormous change in resistance value. I've replaced them with 100K ohm, 1 watt resistors. Brief testing shows promise -- the carrier suppression seems to be a bit more in line, now.)

2. Noticeable distortion on very loud receive signals. Surely AGC related, but I don't yet have a fix...

Notes:

1. Voltage Chart Errors. Take the voltages listed in the manual's "Voltage Chart" with a huge grain of salt. Some are clearly wrong, such as the voltage on V16 pin 6. There is no way it can be 210 volts -- it comes, via a resistor, from the 150 volt regulator!

Similarly, some of the "positive" voltages are actually negative (unless I am really screwing up my measurements!).

Resources:

Yahoo Group: KW-Radios This is a great resource for schematics, manuals, etc.

KW Atlanta Photo: Atlanta

Yaesu FT-1000D AGC Mods

2009-10-15T06:48:00.001-07:00

Some years ago I purchased an FT-1000D. After using it for awhile I began to notice subtle distortion on SSB receive audio that, over time, I found more and more annoying.

The distortion artifacts were subtle but noticeable. They sounded like a slight "crunching" or "crackling" sound (very noticeable when someone says "ahhh."), and could be made to stand-out (when looking for its presence) by rotating the "Shift" knob to accentuate high frequencies. With the shift in its normal position, the distortion was still present, but it tended to be masked (in most cases) by the higher level of the voice signal.

(One quick test I use for locating the cause of audio distortion is to reduce a receiver's RF Gain. If the audio sounds clearer with less RF gain, then, in my experience, there's a very good chance that AGC action is creating the distortion.)

I reduced the FT-1000D's RF Gain. The the audio sounded clearer. Ah ha! There was a very good chance that the distortion, therefore, was related to AGC action, and so I started experimenting...

My main concern was SSB operation (for which I usually use either SLOW or MEDIUM AGC rates). Looking at the schematic, I noticed that there was a 10K resistor (R2140) in series with the 2.2 uF cap used for Slow AGC. Shorting-out this resistor reduces SSB distortion. But it will distort the CW envelope. My feeling was that I never use Slow AGC for CW, so this was an acceptable compromise to make.

Even the 4066 analog switch introduced some distortion artifacts (per its datasheet, its resistance is about 500 ohms for a 10V supply voltage). I replaced this analog switch with a relay, which has a much lower "ON" resistance.

I also replaced the analog switch used for switching in the MEDIUM AGC circuit (the 0.47 uF cap) with a relay. Experiments for FAST and MEDIUM AGC settings revealed that, for CW, a 30K resistor actually worked better than the original 10K ohm resistors used for these time-constants. So I added a 30K resistor (replacing the two 10K resistors) that is common to both the FAST and MEDIUM caps (0.22 and 0.47 uF).

However, the 30K does produce distortion artifacts on SSB. So I decided to compromise: only for strong signals is the 30K ohm resistance switched in -- for normal or low-level signals, the resistance is very low. (For low-level signals, the AGC line sits high. This turns on a 2N2222 which in turn shorts out the 30K resistor. For strong signals, the AGC line is driven lower, until, for very strong signals, it drives the 2N2222 into cut-off (threshold set using the two series-diodes attached to the base), which then places the 30K back into the circuit.)

I found that the 30K also increases distortion on AM signals, so I also short it out for AM operation. (Therefore, it is only present for strong CW, SSB, and, I suppose, FM (it has no effect on FM)).

Here's my original markup of the FT-1000D schematic:

(Click on Image to Enlarge)

Here's a drawing that might be a bit clearer...

(Click on Image to Enlarge)

Notes:

1. If one would like to keep the changes simple and not incorporate all of the mods that I made, I would recommend the following two modifications:

Short out R2140 (10K ohms: Slow AGC.)
If you use Medium AGC for SSB, then also short out R2141 (10K ohms: Medium AGC. But note that this may create some fuzziness on CW signals.)

2. While reviewing my notes from 2003, I discovered a mention in my lab notebook of the addition of a Schottky diode across the 1.5M resistor (cathode connected to the RF GAIN side of the resistor) so that, as RF Gain is turned down, the AGC follows with little delay (otherwise there could be a long delay for the audio level to catch up with the control position. But I don't show this diode in my final schematic. I don't know if it's actually there, or not, and I'm not about to reopen the FT-1000D to find out.

3. I used Clare DSS41A05 Relays.

4. Audio envelope testing was performed by modulating the RF signal from an HP 8640B (in AM mode) with a pulse generator (100ms pulse, rep rate of 1 second, 2 ms rise/fall, 1.4v peak, -1.1v DC offset). The AGC waveform was monitored @ TP2005, and the audio envelope monitored at the headphone jack.

5. Diodes are in series with the base of the transistors to ensure that they fully turn off when the control signals go low.

Important note:
I made these modifications to suit the style of operating that I prefer (SSB ragchewing). These modifications may not be suitable for your style of operation. So, if you're also experiencing annoying distortion, please consider these mods to be a starting-point for your own experiments in improving the performance of the 1000D.

Central Electronics CE 100V Transmitter

2009-10-06T08:48:00.000-07:00

My adventures bringing a CE 100V back to life!

A number of local hams have Central Electronic 100V transmitters, and on occasion I've joined them during their ragchewing roundtables on 80 meters (sans 100V on my part). I enjoyed the sound of the radio as well as its styling, and I thought it might be nice to have one of my own. And thus began my search for a 100V.

I finally found one that a local ham was selling on Ebay. I made a bid...and won it! Fortunately, because the seller was local, I was able to save shipping charges (it is a heavy radio!) and pick it up myself.

When I got it home, I discovered the radio, besides having extensive cosmetic issues, also had operational problems, and I put it to the side while I worked on other projects that were less daunting than tackling the 100V appeared to be.

Finally, I decided to bite the bullet and get the 100V on the air. First thing to do...pull it out of its cabinet and then get to work...

The radio is a marvel of design, and arguably represents the high-water mark of amateur radio transmitter design of the 50's. Which translates into a radio that is large, heavy, and complex (26 tubes!).

(Top View, with VFO Assembly removed)

(Paging Doctor Frankenstein!)

When I looked inside the actual radio, my heart sank...the chassis wasn't dirty, it was oxidized. I believe it must have originally been plated, and this plating had turned an ugly grey color. And in some places, actual rust had appeared!

(Typical oxidation/corrosion on this radio. Labels on the back panel and on the chassis are essentially unreadable.)

(Despite the terrible shape of the chassis, the front panel actually looks pretty good!)

My VFO was very difficult to turn, and felt "lumpy". Per the Tusa notes (well worth a read) on the 100V, I decided to remove the VFO assembly and take a look at what might be going on...

(Note: the VFO assembly is actually fairly easy to remove. You do not need to drop the front panel! Instead, follow the procedure in the Tusa notes (although please note that for step 7, you should unscrew the two bottom mounting posts from the VFO assembly, not from the front panel)).

After I'd removed the VFO assembly, I was curious to know how it looked inside, so I removed the back cover. Whoops! Chunks of foam (and foam bits) tumbled out. Looks like Central Electronics used this foam (3/8 " thick) to act as an insulator to minimize temperature variations within the can. And after 50 years, it was disintegrating.

(Disintegration of the insulation foam in the VFO Assembly!)

I happened to have an old mouse pad lying (1/4" neoprene), so I cut it up and glued it to the inside of the can with some RTV cement. Voila!

(Old mouse pads have many uses, such as...new insulation!)

Now to attack the difficult-to-turn VFO. The Tusa notes recommend repacking the bearings, but from the instructions I'd read (and from the stories I'd heard), it sounded like a real nightmare.

Instead, I decided to see if a shot of WD-40 into the bearings would help to loosen up the old grease...

I held the VFO so that the knob was pointing toward the floor, then applied a quick burst of WD-40 into the "well" (see photo below). The bearings are below this well, and, by holding the knob towards the floor, I hoped the WD-40 would flow down into the bearings.

It seems to have worked. The mechanism turns much more easily now. Sure, I probably ought to repack the bearings (because the viscosity of the grease might give it a bit "smoother" feel). Maybe next year...

(Click on image to enlarge.)

While I had the VFO out of the radio, I was curious to learn how the VFO tracking mechanism worked...

As the VFO frequency is adjusted, the VFO's lead screw moves a core in and out of the main VFO coil, thus changing the oscillator frequency. But there is a secondary adjustable coil, too, whose core is attached (via a rod) to a right-angle bracket that can pivot. This rod moves in and out of the secondary coil according to the height of the "VFO Corrector Adjustment Screws," and allows small corrections to be made to the VFO frequency as the the user tunes over the 1 MHz-wide range of the VFO.

You can get an idea of how the mechanism works from the two photos below:
The lead screw moves the frequency correction assembly (consisting of the screw run through the block) along either direction of the lead screw (depending upon whether the frequency is being adjusted up or down). A rod runs over the bottom of these screws (shown bottom-up in the photo above), which in turn causes the metal right-angle bracket to which it is attached to pivot.

Attached to the other end of this right-angle bracket is a rod which drives the core of the secondary coil in or out, thus correcting the frequency. In the photo below you can see both the larger main coil (on the same axis as the lead screw) and the smaller secondary coil below it (only a couple of turns of this coil are visible).

Pretty clever!

A note about the frequency correction adjustment. I would recommend that you start at the end of the VFO that has the largest "positive" (rather than negative) delta from the dial frequency. In other words: if the offsets at either end of the dial are both positive, start at the end that has the largest positive delta. If both of the offsets are negative, start at the end that is closest to the dial frequency, and if one end is positive and the other negative, start at the positive end.

"Zero" your dial at this frequency by moving the black line on the clear plastic to overlay the "0" on the dial (there's a screw a few inches below the VFO knob that let's you do this). Then, moving the VFO in 500 KHz increments, adjust the frequency using the "VFO Corrector Adjustment Screws" per the Tusa Consulting note on VFO Recalibration.

Problems that I've run into:

Meter not working. No movement, at all. I opened up the meter and discovered that one of the "spiral springs" that attach to the armature had opened up. It was a real pain to repair, but repair it I did. (By the way, my meter is about 1 mA Full Scale, and has a resistance of 47 ohms). I also added a pair of diodes (1n5818) hooked antiparallel across the terminals of the meter (to protect the movement from burning out), as well as a 0.1 uF cap -- if you add the two diodes you must include this cap, otherwise your meter may read low in the "Watts" position.
VFO Sticking/Hard-to-Turn (See discussion above)
VFO Not Tracking (See discussion above)
Lack of the -120V Blocking Bias in STBY mode. A 1uF/200V electrolytic cap that was attached to this line (via a 10 ohm resistor in the power-supply section) was shorted to ground. (Note: neither this cap, nor the 10 ohm resistor, appear in the schematics). These parts were probably added to slow-down the transition between STBY and Transmit. I didn't have a 1uF with a high enough voltage rating in the junkbox, so I instead used a 2 uF cap.
Inability to Null Carrier in Sideband Modes. The meter would remain pegged to the right in Null mode irrespective of any adjustments I made to the two Carrier Balance knobs. I measured the forward-voltage of the four original germanium (CK715) diodes in the modulator plug-in module, and the voltages varied wildly (from 0.234 volts to 0.632 volts). I replaced these with HP 5082-2063 (Schottky?) diodes that I had in one of my parts' bins (their Vf was 0.34 volts, and matched within millivolts for all 4 diodes). Works fine now.
Wattmeter: Reads too low, and cannot adjust far enough. Resistance values had drifted over time, and one of the 100 ohms resistors had drifted to 109 ohms. Replaced with 100 ohms, and now can adjust with the pot (although it's almost at its limit).
RF Ammeter: Reads too high, and cannot adjust far enough. The resistors have apparently drifted. The voltage-divider resistors are in an extremely awkward location, so their replacement is very difficult. Instead, I added 120 ohms in parallel with R147/R148, and that brought the voltage into range to allow correction using the pot.
No X-Axis movement on Monitor Scope. Replaced V21 (6U8A)
Low-frequency Noise in Audio, Eventual Loss of Sideband Suppression. I traced this to a leaky cap in the audio phase shifter module -- one of the symptoms was a high DC voltage at an output (pin 8) of the phase shifter. Cap C130 was leaky (but the leakage couldn't be measured with a DVM) -- I replaced this 4711 pf mica cap (actually measured 4739 pf) with 4731 pf consisting of a 4300 pf mica and a 470 pf mica in parallel. (Update: I've discovered that this problem is discussed in the 100V article by Charlie Talbott, K3ICH, in the August, 1996 issue of Electric Radio, and I've implemented one of his mods, which is to insert a 1uF, 400V cap betweenV7 pin 1 and the phase-shift network PS-2 socket's pins 2 and 6 (to isolate C128 and C130 of the phase-shift network from the B+ voltage on V7's plate)).

With these issues resolved, I've deemed the 100V ready for the air:

The 100V in its operating position!

Although the 100V is now up and running, there are still...

Problems I've yet to resolve:

8 MHz Oscillator cannot be adjusted to be exactly 8.000 000 MHz (it remains too low). Even with the adjustment cap at minimum value.
FSK Adjustment does not span 100-900 Hz. Instead, it only seems to have a range of about 150 Hz.
PA "Idle" Wattage (in SSB Xmit, no voice) should be in the range of 60 watts (per the recommendation of others) -- mine is more around 30 watts. (Central Electronics added adjustment pots for both the driver bias and the PA bias adjustment to their 200V transmitter, but these parts are not in the 100V, and to add them involve more surgery than I'm willing to undertake at the moment).

Other notes and Comments:

The schematics can be inaccurate! I've found additional parts, and I've found parts missing, when comparing the actual circuitry to the schematic.

Tusa Consulting has a number of notes on the CE 100V and 200V transmitters. You can find these notes here.

A previous owner had replaced the two batteries internal to the Speech Limiter module with two AA-size alkaline batteries, and had mounted their holder on the outside of the Speech Limiter's case.

To get at the tubes and adjustments beneath the fan in the audio section, just loosen the transformer screw and tilt the fan bracket up...

Some other useful data...

100V I.F. Mixing Scheme
(Click on image to enlarge)

Some manual copies have impossible-to-read voltage charts. Here are clearer copies (thanks to Jon, K6JEK).

Tube Voltage Chart
(Click on image to enlarge)

RF Voltage Chart
(Click on image to enlarge)

Articles on the 100V in Electric Radio magazine:

"Restoration of the Central Electronics 100V," Dennis Petrich (K0EOO), Electric Radio, Number 20, October 1991.
"Observations on the Central Electronics 100V & 200V," Charlie Talbott (K3ICH), Electric Radio, Number 88, August 1996.

(There may be additional articles in Electric Radio. These are the two that I've found.)

Phase Network Simulation [11 March 10]:

I ran a SPICE simulation on the CE 100V's Phase Network and compared the two outputs. Here's the plot (and the schematic):

(Click on image to enlarge)

The solid line is amplitude, while the dashed line is phase.

Sideband suppression versus phase error can be calculated with this equation:

- 20|tan()| (Where "delta" is the phase error.)

So a phase-error of 1 degree (from the ideal phase-shift of 90 degrees) will result in about 41 dB of sideband-suppression; a phase error of 2 degrees: 35 dB; while a phase-error of 10 degrees will result in only about 20 dB of sideband suppression.

(Note: I haven't included in this simulation the small-signal resistances presented by the grids of the tubes that the phase network output drives (I don't know what they are). For ease of calculation, I've assumed that they're infinite. Also, because I don't have any SPICE models for tubes, I just used a transistor as the driver.)

[LTspiceIV, the program that I used for my simulations, is free, and it can be found here.]

Standard Caveat!

I may have made a mistake in any of the above, so use at your own risk!

PRC-47 Modifications

2009-09-14T08:42:00.000-07:00

[2 December 2010: Added an Addendum to the "Replacement of the Switching Supply Power Transistors" section, below.

8 February 2011: Added another Addendum at the very end of this post.]

The PRC-47 is a Vietnam War era SSB transceiver designed to operate from 2.000 to 11.999 MHz in 1 KHz steps. It is USB only, but can also operate CW or FSK, and it's designed to be powered by either a DC supply (from 24 to 28 VDC) or a 115 VAC, 400 Hz supply.

Transmit power (into a 50 ohm load) is rated at 100 watts PEP (High Power Position) or 20 watts PEP (Low Power Position).

Here are some pictures of my PRC-47:

(Click on Image to Enlarge)

Note the modular construction with plug-in modules...

...And an easily accessible chassis:

Replacement of the Switching Supply Power Transistors:

I've had several PRC-47 transceivers in which the two 2N1653 transistors (used to convert the 24 VDC to 24 V "square-wave" AC) were bad. I replace these two transistors (Q1 and Q2, on the chassis, just under the faceplate) with more modern 2N5884 transistors, which seem to work just fine.

[Addendum, 2 December 2010]

The original transistors used for Q1 and Q2 in the Switching Power Supply are Germanium, and are either 2N1166, 2N1653, or 2N2287 transistors (I have one Tech Manual that specs the 2N1653, and another which specs the 2N2287, and others have told me that the 2N1166 is also used in some units).

Germanium transistors are difficult to find, so, to replace the original "failed" transistors, I chose a more common Silicon PNP transistor. The 2N5884 which I use is not a perfect match for the original transistors, but it's close. Here's a comparison of the specs of the three original transistors (from the fourth edition (1969) of Motorola's "The Semiconductor Data Book") versus the 2N5884 (from ON Semiconductor's website):

It's worth noting that diodes CR1 and CR2 in the Power Oscillator are there to limit the collector-emitter voltage of transistors Q1 and Q2 (the 2N1653 transistors) to 26.5 volts, so, assuming the diodes are still OK, the 2N5884's max rating of 80 volts should provide plenty of headroom. (I haven't made any measurements to verify this, though).

If anyone is concerned about breakdown voltage, you might try experimenting with the MJ15004 transistor -- it's rated to 140 VDC. However, its Ic (continuous) rating is only 20A, versus 25A of the 2N5884, so there's a tradeoff. Personally, I'd go with the 2N5884.

By the way -- the higher Vce(sat) of the 2N5884 (or MJ15004) might result in a lower plate-voltage to the PA tube (because the voltage swing at the primary of T1 will be lower (26.5 VDC - Vce(sat)), and thus lower output power. I haven't verified this, but if you find it to be the case, you can try bumping up your DC input voltage to, say, 28V, to help counteract the swing limitation due to the higher Vce(sat) of the Silicon transistor.

(And whichever transistor you use, please report back with your results!)

PTT Not Working?

If PTT doesn't work on your radio, check the following:

Ensure that CR18 is installed in the AF Amplifier Module (I use a 1N4148).
Ensure that the "PTT" wire (green, in my set) is connected to J2.11 on the chassis (this is the DB-25 jack into which the AF Amplifier Module plugs). If not connected, you might find it tucked to the side.

(Note: It's possible that very early versions of the PRC-47 don't have a PTT function, but are VOX only. I have an "Advance Copy" of TM 11-5820-509-35, dated November 1963, which shows no connection to pin 11, nor the existance of CR18 in the schematics (I presume the set, at that time, was VOX only). However, my later (electronic) version of the manual, dated July 1974, shows these connections.)

LSB Modification:

A radio which only operates USB in the range of 2 to 12 MHz is of limited use to a radio amateur. However, the PRC-47 can be easily converted to LSB operation. To do this, you need to replace the mechanical filter in the IF module (which happens to be a 500 KHz, LSB filter) with a 500 KHz, USB filter. And voila, you'll have LSB.

Here's how I make this modification:

Remove the Amplifier-Modulator module from the radio and remove its covers.
Remove the mechanical filter and replace with a Collins F500-Z4 (or equivalent) filter.
Apply +20V to P4 pin 3 (this will supply power to the module) and the 20V return to the module case. (I find it easiest to attach +20V to the far left side of L9.)
Set a signal generator to 501.5 KHz and apply the signal to J3.
Measuring at J1 (with either a scope or spectrum analyzer), adjust the generator's level so that you see a signal (but don't overdrive the module), then...
peak the measured signal by adjusting the two variable caps, C15 and C17.

AGC Modifications:

In my opinion, the "stock" PRC-47 AGC design results in severe and unnecessary distortion of the receive audio signal. One of my first goals was to attempt to improve the quality of the receive audio. In trying to fix the AGC I'd find a solution to one problem, only to then have another problem (previously hidden) reveal itself to me. And so the modifications, like coral, grew by accretion. Thus, they may not all be necessary (because a later mod may actually cancel the need for an earlier mod), but it would take much more time to determine which mod is irrelevent and which is not, so I've left them as I've implemented them.

First, a bit of background...the PRC-47 AGC is audio derived and results in two AGC signals: -AGC (which controls the gains of the the input RF preamp tubes) and +AGC, which controls the gain of the IF stage.

For SSB operation the PRC-47 has, in my opinion, an AGC decay time which is much too fast. The +AGC line is the dominant actor for normal SSB signals, and thus, to increase the decay time of C42, I changed R74 from 22K to 220K, and then added an emitter-follower (2N2222) to keep C42 from being loaded by successive stages.

I found, though, that when I did this, signals at normal "everyday" signal strengths sounded good, but very strong signals still distorted. A bit more investigation revealed that, with very strong signals, the +AGC signal was being driven so high that it was driving the IF stage transistors (Q2 and Q3 in the Amplifier-Modulator module) into cutoff!

I fixed this by limiting the the level to which the +AGC signal can go with a 13V zener (which, in my radio, results in a clamp voltage around 12.6 volts). Not elegant, but it keeps the IF transistors out of cutoff, and the -AGC signal (which kicks in at higher levels, and which is not limited) performs the AGC function for those very loud signals. (By the way, the 390 ohm resistor that I added between the emitter of the Emitter-Follower and the +AGC line limits current when the zener is driven into conduction, and the 22K provides the original resistance-to-ground as seen by the Amplifier-Modulator module and provides a necessary bias path for the transistors in the Amplifier-Modulator module.)

(I actually use the signal at the emitter of the emitter-follower to serve an "S-Meter" function (described later) because, at the emitter, the AGC voltage isn't clamped by the zener. Thus the S-Meter covers a wider voltage range than it would have otherwise if I'd used the +AGC signal. This signal (from the emitter-follower's emitter) I call "Buffered AGC+", and I connected it to a spare pin on the module's DB-25 plug (pin 12) so that I could then route it over to the meter circuitry located elsewhere in the chassis.)

Another problem I encountered: after releasing PTT, there was increased noise from the speaker (lasting for about a second) until the AGC stabilized the signal level. When I monitored the voltage across C42, I'd see its voltage actually drop momentarily (thus increasing receiver gain) when I transitioned from Xmit to Receive, and then it would recover. Removing C35 eliminated this noise burst.

After I had removed C35, I discovered that sometimes I'd lose output power during xmit. When this occurred, I noticed that the +AGC voltage was rising (and thus cutting off IF amplifier gain). I suspected that noise on the +26VDC line might have been affecting AGC during transmit (because C35 was removed), and I modified the circuit to use the +20 VDC line instead. Note that this required paralleling R59 with a 33K resistor to keep the junction of R59/R55 at around 8 - 9 V during transmit.

Essentially, during transmit relay K1 applies 26 VDC to the VOX line. This turns on a 2N3904, which in turn switches on a 2N3905 which connects +20 VDC to R59 (less, of course, a Vce(sat) voltage drop), rather than the original 26 VDC.

Another issue: during transmit, because of the newly-added 2N2222 emitter follower, the voltage across C42 can drop to near 0 volts. Then, when switching back to receive, it's possible to have a "pop" on the attack of loud signals because this low voltage causes the gain to be too high.

I fixed this by clamping the voltage across C42 during transmit: a 2N3904 transistor turns on (during transmit), which forces the voltage across C42 to be about 3.8V (the 3.8V comes from a 3.9V zener). Then, when we transition back to receive, there's less of a difference between initial receive gain and the required receive gain. (We could actually have used a zener with a bit higher voltage (nearer, say, 4.8 to 5.4 volts), but 3.9 volts seems to work fine.)

Here are some voltage measurements on my PRC-47 with these mods:

Receive, no antenna connected: V(C42) = 0.4Vdc V(Buffered AGC+) = 5.4Vdc
Receive, 80 meters, w/antenna and atmospheric noise: V(C42) = 9.4 Vdc V(Buffered AGC+) = 8.9Vdc
Transmit: V(C42) = 3.8 Vdc V(Buffered AGC+) = 5.4Vdc

Here's a schematic showing these AGC Mods.

(Click on Image to Enlarge Schematic)

Here's how the implementation looks:

Adding an S-Meter:

Normally, the PRC-47's meter doesn't move when the radio is in Receive mode. I thought it might be nice to have some sort of indication of relative signal strength (having a "dead" meter always seems a bit unnatural to me). If you don't mind a meter that's uncalibrated and non-linear, then here's a simple mod you can make.

It does require using a spare pin on the AF Amplifier module to run a "Buffered AGC+" signal to the outside world (and a wire added from the DB-25 jack (J2 on the chassis) to the new components mounted elsewhere on the chassis (see photo below for component location and mounting -- I used pin 12 of the DB-25 (J2) for this new signal). You can find a description of this "Buffered AGC+" signal above, in my AGC mods. (A "buffered" agc signal is used to drive the meter in order to keep the meter from loading the AGC cap: additional loading would worsen agc performance by shortening the agc decay time.)

I simply "diode-OR'd" this new AGC voltage with the existing "Xmit Signal Strength" signal. "Diode-ORing" simply means that whichever of these two signals has the highest voltage level will be the signal which controls the meter reading.

During Receive, the "Buffered AGC+" signal runs from about 5.4V (no antenna attached) to around 17.7 volts max (for strong signals). And during xmit this signal is at 5.4 volts. The 7.5 volt zener keeps the meter at 0 when there's no antenna attached (because 7.5 volts is greater than 5.4 volts, the zener doesn't conduct), and there's a couple of extra volts of head-room to keep the needle at a reasonable (left-side of meter) deflection when receiving normal atmospheric noise. And because this zener doesn't conduct during TX either, only the Xmit Signal Strength signal feeds the meter during TX.

Xmit Signal Strength is actually a fairly low level signal (if I recall, from my measurements it's about 250 mV max), and thus I used a Germanium diode (1N34) for this side of the diode-OR because of this diode's low turn-on voltage.

The 200K resistor limits the current to the 50 uA meter so that, at the strongest receive signal levels, the meter is just at its maximum deflection.

Here's the schematic:

(Click on Image to Enlarge Schematic)

And here's its implementation. I added a terminal strip for the additional components.

Transmit Audio issues:

The PRC-47 transmit audio leaves a LOT to be desired. It has an audio "compressor" which will distort the signal. Carbon mics can sound crummy. An inordinate amount of crud from the switching supply is coupled into the signal, manifesting itself as an underlying "whine" sound.

The conversion of the 24 - 28 Volt DC source to AC (actually, a square wave), which is required to generate the other voltages that the radio requires, unfortunately results in nasty switching artifacts that appear everywhere throughout the radio (on the +20VDC line (and other signals)). The ultimate result? Whine on the TX audio.

I've worked for many frustrating hours trying to minimize this whine. I've had some success, but I've never been satisfied. Because I use an "amplified"D-104 as my mic, I finally resorted to a simple solution: apply even more gain to the D-104 signal, EXTERNALLY (so that it's much louder than the internally-coupled noise), and bypass the radio's internal gain stages (into which the noise was being injected and amplified). This doesn't completely eliminate the noise, but the noise does seem to be much less and just about tolerable.

During my attempts to reduce this whine, I tried any number of fixes. Some seemed to improve things, such as the addition of 0.01 uF from the "CW Key" signal to ground (I mounted this on the back side of the middle board in the AF Amplifier module). But many of the mods I tried had little effect.

(One likely area of coupling might be the "Mike Input" wire. This wire runs from the Mic Connectors on the front panel to J2 pin 25 on the chassis (the AF Amplifier module's connector), and during its run from the front panel to J2.25 it's bundled together with quite a few noisy wires in a wiring harness. Thus, noise coupling onto this wire is certainly a very real possibility. One mod I'd like to try to reduce this possible coupling is to replace the wire with a shielded wire (e.g. coax). Unfortunately, on my radio, the Mic Connector pins to which this wire connects are difficult to get to. So, I'm leaving this for another day...)

Here's the External Mic Preamp which I built to drive the PRC-47 (I needed more gain, even though I'm using an "amplified" version of the D-104 mic):

And here are the mods I made to the mic amplifier side of the AF Amplifier module:

(Click on Image to Enlarge Schematic)

(I recognize that this solution really only suits my particular situation, and it is of little use if one wants to use a carbon handset or mic with the radio. So I encourage readers to experiment and discover what works for them. One tip I can give -- a friend, Dick (W1QG), replaced R2 (47 ohms) with a current source (about 600 uA), and also added emitter degeneration to Q1 and Q3, to lower the gain of both of these stages. This apparently helped out when using a carbon mic.)

A note on how I adjust the "MIC AMPL GAIN" pot (R27) on the AF Amplifier module...

Adjustment of this pot is described in section 3-26 of the Tech Manual (TM 11-5820-509-35). I'm lazy, so rather than attach a generator to the mic input, I adjust R27 using the TUNE signal (when in Tune Mode) so that it measures 3.5 vpp at the measurement point (see section 3-26.b). Then, once I have R27 set to 3.5vpp for the TUNE signal, I adjust the gain of my EXTERNAL mic preamp to also give 3.5 vpp at the same measuring point (for voice peaks). (Note, diodes CR5 and CR6 act as limiters, and they limit both the TUNE and the mic's audio signals -- if you're using an external mic preamp and you set its gain too high, you'll grossly clip your audio signal, which can add distortion. The TUNE signal, by the way, isn't a nice sine-wave, but is clipped by CR5 and CR6.)

If the radio is already buttoned up, I adjust the gain of the External Mic Preamp to give the same envelope peak voltage on the RF output (monitored with a 'scope) that I get in the TUNE position.

Curing that "Donald Duck" sound on Sidetone:

During Transmit the PRC-47 can insert a small amount of the TX audio back into the receive path so that operators, while transmitting, can hear themselves talking. This audio is called "sidetone", and its usage comes from the telephone world, where it was found (allegedly), that if people heard themselves in their telephone handset's earpiece while they talked, there was less of a tendency for them to yell into the mouthpiece.

Unfortunately, the PRC-47's sidetone signal can sound very distorted (I liken it to "Donald Duck"). The cause seems to be an audio envelope that appears on the "Sidetone Gate" signal during transmit. I improved this by adding a 4.7 uF cap from the "Sidetone Gate" line to ground in the AF Amplifier module (cap '+' goes to ground, cap '-' goes to Sidetone Gate).

If using an external speaker in lieu of a handset (which I do), sidetone can be quite annoying, so I also turn the sidetone gain potentiometer (R46 on the AF Amplifier module) all the way down.

Other Notes:

1. Extender Cables.

When working on the PRC-47, it's nice to have a set of extender cables that will let you remove modules from the chassis for debugging/experimentation, yet allow them to remain attached. Cables for this purpose actually exist, but are difficult to find. However, it's easy enough to make one for the AF Amplifier module (which uses a DB-25 connector). Here's the one I made (shown in use):

2. Documentation:

Get the latest version of the Tech Manual (TM 11-5820-509-35). The one I have is in electronic format (PDF), dated July 1974. Various modules have been modified over the years (from the initial design), and it's worthwhile having the schematics for the latest designs (Chapter 1 lists the various changes that have been made to the radio since is inception).

However, there is one caveat regarding this manual: there ARE errors! The text was (apparently) read in via OCR, and the usual goofy OCR mistakes are the result (I guess proof-reading was either sloppy or non-existant). Also, the schematics can have errors. I've run across a few, but, fortunately, not many.

(Also -- I always keep a lab notebook nearby in which I jot down notes, modifications, and measurements as I'm going along. It's a habit I got into as an engineer and it helps tremendously when, years later, you're looking over a schematic and wondering, "Why the hell did I do that?" Which is exactly what happened to me when I started writing up this post on the PRC-47 -- many of these mods were made six years ago, and it was no longer clear to me why I'd made some of them. Fortunately, a quick glance through my lab notebook quickly resolved any questions that I had.)

3. Pot Settings.

I usually set the following pots as follows:

On the AF Module:

R46 (Sidetone Gain): Full CCW
R52 (AGC Gain): Full CW
R54 (Rcvr Gain): Set per manual (section 3-22). Or use your ear.
R27 (Mic Ampl Gain): See Note above (in the "Transmit Audio Issues" section).

4. Setting TX Gain:

See sections 3-27 and 5-4 in the Tech Manual.

5. Power Amplifier Bias Adjustments:

Per Dick, W1QG, the PA bias should be set so that the PRC-47 "idle" current (from the 24 volt DC supply) during transmit is 6 amps ("idle" means that there is no TX audio). Using the instructions in section 3-27.c of the Tech Manual, adjusting the bias so that the voltage at A5J2 is -140 volts, instead of -110 volts, seems to achieve this goal.

6. Replacing bad transistors in modules...

Many (if not all) of the transistors used in the PRC-47 are germanium, not silicon, devices, and, if they go bad, identical replacements can be difficult to find. Whenever I come across a bad transistor, I've had good success simply replacing it with a silicon device. For example, I might replace a bad PNP with a 2N3905, and a bad NPN with a 2N3904 or 2N2222 (because I have lots of these devices lying around my "lab"). Silicon devices have a higher Vbe, but not significantly higher considering that much of the PRC-47's circuitry is biased from 20 volts. Thus, I don't believe that use of silicon devices will significantly alter the bias point (and therefore, potentially, the gain) in much of the PRC-47 circuitry.

Also -- pay attention to the transistor's application. You want to ensure that whatever replacement transistor you choose won't get smoked! Fortunately, you can still locate specs for many of the original 'Ge' transistors on the web, and a quick comparison with a replacement 'Si' transistor's specs should tell you if your choice is adequate.

A Final Note!

It's always possible that I've made a mistake in these notes (or during the implementation of these mods). If there's something that looks wrong or suspicious to you, please feel free to contact me and let me know.

Many thanks!!!

- Jeff, K6JCA

Addendum, 8 February 2011 --

I've just received a note from Ron Boltz, K3TZJ, who writes:

Finished my PRC-47 sets that were here with two more arriving this week. I sent you some other info a month or so ago. Two sets had 2N2638 transistors in them so another number to add to the list.
I got to do some testing on the last set with 2N5884’s installed. The frequency of the inverter goes up to 525Hz and the high voltage goes up to 1,710. The bias also increases to -132 volts. I discovered that the bias needs to be adjusted on some sets as the PA tube overloads if not set correctly. Several sets had the bias way low for some reason.

Three of the sets I worked on had the key lines cut at two places. Drove me nuts till I discovered them. One cut prevented the 800Hz oscillator from running during the “tune” cycle and the other place was on the PA over temp switch which prevented keying. All three sets were Marine Corp sets and had the same depot stickers on them. I wonder if this was a method of de-mil?

Ron Boltz

K3TZJ

http://www.rattrig.com