Quickie Pneumatic Antenna Launcher

[21 Feb 2015 Update:  for an improved design, please see this newer post:  Improved Antenna Launcher.]

I need to get wire-antenna supports up into some tall pines at a remote location, and the slingshot that I would normally use to do this is at my brother's house. So...in its absence I thought I'd instead make a "pneumatic antenna launcher" to help me get the supports up into high tree branches.

A quick Google search revealed a number of plans for pneumatic antenna launchers, the most common using 2.5" PVC pipe. Although these designs were usually pretty fancy (using adapted sprinkler valves to trigger the launchers), I thought they might form the basis of a simpler design that I could quickly assemble. So off to Home Depot I went to pick up some 2.5" PVC and accessories.

Unfortunately, when I arrived I discovered that the local Home Depot only has Schedule 40 PVC pipe up to 2" inner-diameter, but not 2.5" pipe.

Well, why not use 2" pipe? With this diameter in mind, I searched through the bins of various PVC couplings and parts, designing the launcher in my head as I discovered what bits and pieces Home Depot had in stock.

With money dispensed, home I went, and not much later I had my launcher! Here it is:

(Click on image to enlarge)

The air-chamber and barrel are made from 2" I.D. Schedule 40 PVC pipe. Overall length is 90 inches. The barrel is 32 inches long, and the air-chamber is 52 inches long (roughly 2.5 quarts in volume).

I chose 2.5 quarts as a compromise between air-volume and length of the chamber. Other designs that I found on the internet seemed to use a volume of about 3 quarts for their air chambers, but, with 2" PVC pipe, this would require a chamber length of 60 inches, which I thought would make the overall launcher a bit too unwieldy. So I shortened it up a bit, which, for me, puts the "trigger" at a nice height when the end of the launcher is resting on the ground.

For the "trigger," rather than try adapting an expensive sprinkler valve as others had done, I went with a low-tech, low-cost ball valve which I'd seen used in the following photo of a potato launcher.

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James and Devin with potato launcher (circa 1999?)

I chose a 1/2" ball-valve after I discovered, while testing various size valves at Home Depot, that it was the one that I could turn the easiest:

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The 1/2" ball-valve is threaded at both ends. To connect it to both the 2" air-chamber and the barrel, I screwed into each end of the valve 1/2" (threaded) to 3/4" (female slip) adapters (with a generous amount of Teflon pipe-tape on the threads), and then I glued short lengths of 3/4" PVC pipe into the slip-joint ends of these adapters. In turn the other ends of these short lengths of 3/4" pipe are glued into 3/4" (slip) to 2" adapters. The barrel and the air-chamber connect to these 2" adapters via 2" slip couplings (again, glued).

Note that the threaded couplings allow the launcher to be disassembled for easier transport. And, should I ever decide to change to a fancier trigger mechanism, they would allow me to easily swap out the original ball-valve trigger for something different.

To fill the air-chamber I used a Presta valve from an old bicycle inner-tube that I had lying around. It's threaded and has a nut, which eases its installation.

(Click on image to enlarge)

A Schrader valve would have been preferred, as Presta valves are a bit fragile, but the Presta valve was what I had on hand.

To ensure a good seal between the valve and the air-chamber pipe, I cut out two pieces of the bicycle inner-tube rubber, each piece roughly a circle 1" in diameter. Into the center of each piece of rubber I cut a small hole slightly smaller than the diameter of the Presta valve. I pressed these each over the valve and worked them, one at a time, down the stem to the end that would be within the air-chamber pipe.

I drilled a small hole in the pipe just past the point where the end-cap would stop (do NOT attach the end-cap yet before you install the valve!), and then I inserted the valve into this hole. With its nut tightened down, the rubber "gaskets" I'd made provided a good seal against the inside of the air-chamber.

After I'd installed the valve, the air-chamber was capped off with a 2" PVC cap, glued in place.

Because the pipe is only 2" in diameter, I couldn't use normal size tennis balls. A visit to Jon, K6JEK, and his wife revealed exactly what I needed. Their dog Buster likes to chase 2" tennis balls.

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I tested one of these tennis balls in the launcher, and it worked great! Buster was too attached to his tennis ball for me to try to take it (and his others were chewed beyond recognition), so it was off to the local Petco (pet supply) store to search for more 2" tennis balls!

(Click on image to enlarge)

The yellow balls are a bit softer than the blue/pink ball, and they are "squeaky" toys. I drilled a couple of holes in one so that I could insert a tie-wrap to use as an attachment loop. Then, at the other end, I cut a thin slit with an X-acto knife so that I could insert pennies to add weight. Per another website, the ball should weigh between 4 and 5 ounces (as the best tradeoff of height, safety, and the ability to pull the line down over tree branches and foliage). Getting it up to 5 ounces pretty much fills up a 2" tennis ball with pennies! (Each penny is roughly 0.1 ounces).

Here's a finished tennis ball, with tie-wrap attachment loop:

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Using a bicycle tire-pump, I've tested my chamber up to about 80 psi and it seemed to hold its pressure fine (at least for the time it took me to insert a ball and launch it). The 2" pipe itself is rated to 280 psi (and the 3/4" pipe to 480 psi), but the ball-valve is only rated to 150 psi. I'd recommend keeping the max pressure well below this point, though.

A small paint bucket can be used to hold the line and keep it from becoming entangled in ground debris (e.g. twigs and leaves). Tie one end of the line to the bucket handle!

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Results:

Shooting the weighted 5 oz. tennis ball straight up into the air resulted in the following heights:

• 20 psi: 15 feet
• 40 psi: 35 feet
• 60 psi: 65 feet

(Note: I only tested once at each psi level. Heights are approximate, based on a rough measure of how much line played out).

While erecting my 80 meter full-wave loop, I discovered that I needed the ball to be heavy so that, if it were in an environment with many branches, it had a better chance of pulling down the line attached to it. I had started with a 4 oz. tennis-ball load, but finally decided I was better off with the ball loaded with as many pennies as I could fit into it. The result is a ball which weighs about 5.5 oz.

Even at this weight, sometimes the ball wouldn't drop all the way to the ground, and I would have to "finesse" it down by wiggling the line or trying other tricks. And sometimes I just had to pull the ball back and start over again. Perhaps a more "slippery" line might have helped the ball descend, but in the end I was able to get all of the supports up and the loop raised without either having the ball become permanently stuck in a tree, or my having to run to the store to purchase yet one more thing.

Ready, aim...

Notes:

1. Mechanically, the weakest point is the smaller-diameter pipes and adapters that make up the trigger mechanism: this is where you'll see the launcher bending. To protect these parts when transporting or storing the launcher, I'd recommend unscrewing the barrel from the ball-valve, and not unscrewing the air-chamber. Keep the air-chamber screwed into the ball-valve, because it's important to maintain a good air-tight seal at the threads to prevent pressure loss.

2. More height-per-psi might be achievable with a better (faster) trigger mechanism (e.g. adapted sprinkler valve), but I'm satisfied with my results -- they work for my application, and the design is very simple and easy to construct. Also, because the tennis ball is narrower than 2", air can escape around it as it's moving through the barrel. Some sort of circular disk to minimize escaped air (say, made out of an old mouse pad?) first placed at the bottom of the barrel with the ball then inserted so that it's lying on top of it might improve performance. But in the end I've decided that all I really need to do is add a few more psi with my bike pump to get the heights I need.


Resources:

1. Here (An excellent site!)
2. 2" ID launcher

(Googling "spud gun," "potato gun, and "tennis ball launcher" will provide other sites with great ideas, too.)


Caveats:

If you build one of these, use common sense and, above all, use at your own risk! Follow instructions for gluing PVC, allow adequate curing time, and, when finished, don't overstress the PVC by pumping in too much air!

Solid-stating the Heathkit HR-10 Receiver

In a previous post I detailed my experiences in modifying a Heathkit HR-10B receiver. Although performance improved, I've never been entirely satisfied with those mods, mainly because, to my ears, there is a very subtle distortion that seems to occur with loud signals. I suspect that the input into the NE602 product detector stage is a bit too high (because the AVC isn't doing a great job limiting signal levels?), and the oscillator is being pulled slightly on high-level signal voice-peaks.

Rather than continue to incrementally modify that HR-10B receiver to improve its performance, I thought an interesting project would be to completely solid-state an HR-10 ( or HR-10B). However, I didn't want to rip the guts out of the HR-10B that I was currently using -- it was in too nice a condition, physically, for that (which is why my mods that I'd made to it can be easily backed-out).

Luckily, I found a junker HR-10 receiver that was exactly what I was looking for: rusty, almost complete, inexpensive, and looking for a home -- the perfect playground for experimentation!

Here's the top of the chassis, as received. Just a wee bit of oxidation...

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And here's the bottom of the chassis. Everything looks like its there!

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I started by removing all of the parts except those I expected to use (e.g. the RF transformers, Oscillator tank components, and the variable caps), sanded the chassis to remove the oxidation, and then I began designing, building, and testing...

The final receiver chassis: rust removed (via sanding), modifications installed, and ready to receive signals!

(Click on image to enlarge)

Schematics:

Here are the schematics for the new receiver:

Page 1: RF Input

This page contains the Input RF Filters, the first Mixer, and its VFO. It uses the original RF bandpass components (L1-L5 and their associated capacitors) as well as the Oscillator tank components (L11-L15 and their associated capacitors).

(Click on image to enlarge)

Notes on Page 1:

• All components with reference designator values less than 100 are original HR-10 components.
• Replaced the antenna connector with a BNC.
• I couldn't get good performance using the NE602's internal oscillator with the existing HR-10 oscillator tank circuits, so I designed a separate oscillator using a J310.
• The 8.2 ohm resistor and ferrite bead the Q101's gate ostensibly prevent VHF oscillations, but I've not verified if they really do any good, or not.

Page 2: IF Filter, IF Amplifier, and AVC

This page contains the IF Filter, the IF Amplifier, and the AVC circuitry.

(Click on image to enlarge)

Notes on Page 2:

• This page has been updated [13 Sept 15] to Rev 2.  See the comments at end of this blog post regarding revisions to this page.
• All components with reference designator values less than 100 are original HR-10 components.
• There is roughly 20 dB of loss through T1 and the crystal filter, which Q201 compensates for (plus a few dB).
• The original loads for the MC1350 had been just the 330 uH inductors, but the circuitry was unstable. Adding 5.1K resistors in parallel with each inductor calmed it down. I didn't bother to try it with just the 5.1K resistors as loads.
• D201 prevents the AGC (aka AVC) voltage that drives the MC1350's AVC control pin from exceeding the MC1350's power supply.
• One output of the MC1350 drives the SSB demodulator (single-ended). The other output is used to derive AGC from the 1.68 MHz IF signal. Thus AGC is IF-derived, not audio-derived.
• But first the IF signal is amplified by Q202 and Q203 before it is rectified by D202. (This amplified signal will also be used as the source for AM demodulation on page 3).
• Similar AGC voltage results were achieved whether D202 was a silicon, Schottky, or germanium diode. So I left the diode as a silicon one.
• C217 provides an RF "ground" for the AGC reference rail (U202.8), and R224 helps isolate the op-amp's output from any high-frequency IF signal (or rectified IF signal) that might appear on this rail (via the AVC cap, for example).
• The input of the MC1350, when driven single-ended, cannot exceed 2.5 Vpp or else distortion occurs at its output.
• As the output of the MC1350 driving the NE602 SSB demodulator (on page 4) is increased from about 20 mVpp, the audio signal becomes more and more distorted (although this distortion might be difficult to hear). For example, given a signal generator's signal that's been tuned in by the receiver so that it produces a 1 KHz audio signal at the speaker, if the level of the IF signal from the MC1350 is 30 mVpp, the audio second harmonic (2 KHz) is about 40 dB down from the fundamental. If the MC1350 output is increased to about 100 mVpp (by reducing AGC loop gain), the second harmonic increases to be only 20 dB down, and there is noticeable "pulling" of the BFO oscillator frequency. For this reason I set the AGC loop gain (via R228) to keep the MC1350 output level at about 30 mVpp so that the second harmonic was 40 dB down from the fundamental. There is some IF signal overshoot (to about 50 mVpp) when a -30 dBm signal goes suddenly from OFF to ON, but there is no noticeable audible "popping" at the speaker from the overshoot. (Note: overshoot worsens as R228 (AGC Loop Gain) is decreased in value -- this also corresponds to an increase in output level from the MC1350 and increased harmonic distortion at the demodulated audio output, as previously discussed.)
• With D202 a silicon diode and the gains set by the component values shown in the schematic, AGC action doesn't start to limit a signal (on 80 meters) until the input signal level reaches about -100 to - 90 dBm. From that point, the AGC Voltage (at pin 5 of the MC1350) increases in 0.03 volts steps (roughly) for each 10 dB step in input signal level until the input signal reaches about -30 dBm, at which point the input stage (NE602) limits the signal. AGC Voltage varies from about 3.92 volts (no signal) to 4.12 volts (input limiting).

Page 3: Demodulation and AF Amplification

This page contains the SSB and AM demodulators and the AF Amplification chain.

(Click on image to enlarge)

Notes on Page 3:

• All components with reference designator values less than 100 are original HR-10 components.
• Q301, when ON, connects the BFO tank to ground so that the BFO can oscillate. R301 provides a DC path for the Q301's collector (as there is no DC path through T5) to ensure that the transistor is always ON.
• In SSB mode, C306 provides a pole at about 5 KHz (with the NE602's output resistance of 1.5 Kohms). And for both SSB and AM, C308 provides an additional pole at about 8 KHz.
• Q302 provides additional amplification of the IF signal so that it can drive the AM detector consisting of diodes D301 and D302 (the lower this signal is, the more clipping occurs on the "low" side of the modulation envelope because the signal doesn't exceed the diode turn-on thresholds).
• C318 compensates for the crystal filter's passband shape (which results in a low-frequency "hump" in the audio when operating AM). Adding a zero at about 1 KHz (C318 = 1N, R318 = 150K) reduces this hump, thus flattening the AM passband so that it sounds less bassy.
• C320 adds a pole at about 4 KHz when in AM mode, helping to reduce the "hiss" of the wideband noise from the AM Detector (detecting noise from the MC1350 output).
• The LM1875 came out of my junkbox. Other audio amps should work fine, too.

Page 4: S-Meter and Calibrator
This page contains the S-Meter and 100 KHz calibrator circuitry.

(Click on image to enlarge)

Notes on page 4:

• All components with reference designator values less than 100 are original HR-10 components.
• The S-Meter amplifier has a gain of about 16, which was a compromise. Because of the AGC control-voltage characteristics which are used to drive this meter (see discussion for page 2 of the schematics), if the gain was set so that the needle was at S9 for a -73 dBm signal, then, as the signal level was increased by 10 dB, the needle would move by 20 dB on the S-meter scale and it would quickly peg on the right side. Conversely, if the gain was set so that the needle for an S-9 + 60 dB signal was at the far right meter tick and the needle moved by 20 dB for a 20 dB change in signal level, the needle sat at about S3 when there was no signal. The problem is that the AGC voltage operates over a smaller range of signal levels than those represented by the meter scale (in which S0 is -127 dBm, S9 is -73 dBm, and S9+60 dB is -13 dBm). So I threw up my hands and compromised with the values shown.
• The 7490 Decade Divider in the calibrator circuit is NOT wired to produce a 100 KHz square wave. Rather, it's wired to generate a 100 KHz signal whose duty-cycle is 20% so that even harmonics, as well as odd harmonics, are produced (a square-wave with a duty-cycle of 50% produces no even harmonics!).

Page 5: Power Supply

This page contains the power-supply and dial-light circuitry.

(Click on image to enlarge)

Notes on page 5:

• All components with reference designator values less than 100 are original HR-10 components.
• The power supplies should be self-explanatory. To remove 120 Hz hum (and its harmonics) from the 17V rail (for low-noise applications), I used a simple filter consisting of R501 and C506.
• The 1815 bulbs are rated at 200 mA each for 14 Volts. Lifetime is 3K hours.
• I use a string of 8 diodes (rather than a resistor) to drop 17 VDC down to something lower for the lamps -- diodes will keep the lamp voltage constant even when bulbs with different current draws (e.g. 1813 or 756) are used in lieu of the 1815 bulbs. Fewer diodes can be used, but the 8 diodes in series give me a brightness I was satisfied with. The voltage across the bulbs is dropped to about 11 volts, and this lower voltage should increase bulb life. At 11 volts the two lamps, together, draw about 0.32 Amps total (measured through R510), which means that each diode dissipates about a quarter-watt each (or 2 watts, total).

Construction:

After removing most of the original HR-10 components, I started building up my new circuitry on sheets of PCB material that I'd screwed to the chassis:

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I used my own construction technique, which is simply mounting components so that I can read their values. ICs are mounted facing up, and I usually fold out their pins (so that they look like wings) and mount them by soldering a couple of the pins of each IC to components mounted vertically on the PCB (e.g. a bypass cap (on the power pin) or to a 1 Megohm resistor that is standing up with one end tacked to the PCB copper plane (you need to first ensure, though, that 1 Meg to ground will not affect the signals using that pin!)).

Other 1 Meg resistors (I have a large reel of them here) are soldered vertically (one end to the copper sheet) to serve as mounting posts for other components. In my opinion, this method is easy and it beats trying to solder or glue little pads made of PCB material to the copper sheet (one technique used by others).

A closeup of my construction technique:

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Yes, I know. It isn't pretty. But it works.



Additional Notes and thoughts:

1. SSB versus AM passbands

With the SSB passband adjusted (via T1) to be fairly flat, the passband in AM mode was very narrow -- noticeably less than 2 KHz, and thus AM signals sounds pretty bassy.

In order to get a bit more frequency range in AM mode, I readjusted T1 to make the AM frequency response fairly flat out to about 2 KHz. However, this put a 7 to 8 dB hump (in LSB mode) at the high end of the audio spectrum.

The plots below show this. The grey graph is the frequency response to noise in AM mode, while the blue graph is the frequency response to noise in LSB mode. (Noise fed to the antenna connector from an external RF noise generator).

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Yes, the hump looks terrible, but during listening tests I didn't find it to be too objectionable on LSB, and so, for the moment, I've decided to keep these passbands as they are (as a compromise between AM and SSB), but I might change my mind in the future. (Note, too, that this hump appears as a bass hump in USB mode).

2. MDS Levels, by band:

By ear (rather than quantitatively), MDS (Minimum Discernible Signal) on the different bands is roughly the following:

80 meters: -130 dBm
40 meters: -130 dBm
20 meters: -120 dBm
15 meters: -90 dBm
10 meters: -110 dBm

As you can see, both 15 and 10 meters are pretty deaf. I've not yet found a solution for this problem, and, because I don't spend any time on these bands, this is not a very high priority for me. However, there does seems to be a bit of VFO blow-by on these two bands which is getting into the AGC detector (and thus adding attenuation to the signal path), which isn't helping. I added a shield between the VFO coil assembly and the MC1350 (consisting of a piece of copper-clad PCB material mounted vertically and soldered to ground) which seems to help reduce this VFO-pickup, but it hasn't cured the problem when 15 meters is selected.

Also, both 15 and 10 meters hetrodyne the signal using the second harmonic of the VFO. If the VFO is clean (i.e. it looks like a sine wave) there will be very little harmonic content and this could affect the conversion gain. Unfortunately, the conversion gain of the NE602 is directly related to the VFO signal level (up to a point), and therefore, in the case of 15 and 10 meters, if the amplitude of the VFO's second harmonic is low, so will be the resultant IF signal, which is why it can sound deaf on those bands (I verified this, by the way, using an external generator as a VFO. With its frequency set to the original VFO's second harmonic (e.g. 22.78 MHz to receive 21.1 MHz)-- sensitivity on 10 and 15 meters improved at VFO amplitude levels comparable to those used for 80 and 40 meters.)

One possible solution might be to add a frequency doubler to the output of the VFO for 15 and 10 meters to increase the amplitude of the second harmonic. We'll see...

3. VFO Frequencies, per band:

80 meters: 5.18 - 5.68 MHz (VFO = F + 1.68 MHz)
40 meters: 8.68 - 8.98 MHz (VFO = F+ 1.68 MHz)
20 meters: 15.68 - 16.08 MHz (VFO = F+ 1.68 MHz)
15 meters: 11.34 - 11.565 MHz (VFO = (F+ 1.68 MHz) / 2)
10 meters: 14.84 - 15.69 MHz (VFO = (F+ 1.68 MHz) / 2)

4. Image Rejection:

On 80 meters I can sometimes hear 40-meter shortwave broadcast stations. For example, if the VFO is tuned to 5.53 MHz (to receive a 3.85 MHz signal), the receiver will also pick up a signal at about 7.21 MHz, which is the image of the 3.85 MHz signal (5.53 MHz + 1.58 MHz). A -70 dBm signal at 7.21 MHz is only about 20 dB down from a -70 dBm signal at 3.85 MHz -- not very good image rejection. An external antenna tuner (low pass topology) can improve this rejection, though.

One way to improve image rejection might be to add an RF preamp prior to the NE602 and use the existing resonant L/C circuits from the original HR-10 RF Preamp (L6-L10). However, some amount of attenuation would probably need to be added, too, so that NE602 isn't overdriven by loud signals. Currently, on 80 meters, it starts to conk out at around -30 dBm, and for that reason I wouldn't want to add additional gain prior to the input NE602, unless this gain is counterbalanced with an equivalent loss.

5. Oscillator Drifts:

There is some amount of drift when the receiver is first turned on, but it seems to stabilize fairly quickly.

On 80 meters, overall receiver drift, from a power-off state, was measured to be roughly 1000 Hz in the first minute. Three minutes later it had drifted another 400 Hz, and from then on it settled down to an overall drift on the order of +/- 50 Hz over an hour.

To separate out VFO drift from BFO drift, when the VFO was replaced by a Fluke 6060A signal generator, the BFO drift, from a power-off state, was measured to be about 300 Hz over one hour.

6. ANL Switch: I haven't yet wired up the ANL switch (nor designed ANL circuitry) -- it's a feature I rarely use, and in the future I might decide to assign a different function to this switch (e.g. selectable input attenuation, or...?).

7. Other oddities:

• On 10 meters you can pick up the 17th harmonic (loud!) of the BFO (at around 28.56 MHz).

Future Improvements:

Someday...

1. Improve performance on 10 and 15 meters (possibly by adding a frequency-doubler circuit to the VFO for these two bands?).

2. Add an ANL circuitry (for the existing ANL switch).

3. More RF filtering to improve image rejection.

4. Add REC/STBY function to octal connector on rear of chassis so that can mute receiver if used with a transmitter.


Caveats:

1. I could have easily have made a mistake, so please regard (and use) this design accordingly.

2. I make no claims that component values are the optimum ones which could be used -- rather, I used values and components which, from the data-sheets and my design equations, seemed to be appropriate choices, and I modified these as needed. The values I've used work for me, but I've not spent any time evaluating the design from the perspective of "optimal" (rather than "good enough") component selection.

Class E/F Exciter for the 813 AM Transmitter

This exciter replaces the Johnson Ranger that I'd originally used to drive my 813 AM Transmitter (described here, here, and here). It uses a modern Class E/F PA (described in further detail here), and it has a separate audio amplifier to drive the modulator deck in the 813 rig.

   

(Click on image to enlarge)

There is one major difference, however, between my original Class E/F PA, which was designed to generate 40 watts of RF power, and this final PA. This difference commences with a big...

   

Oops!

For when I connected this exciter to the 813 rig and keyed it for its initial "smoke test," the 813 Transmitter's grid-current meter pegged. 

Oops! 

But this shouldn't be! I'd measured the power output of the Ranger when it was driving the 813 Transmitter, and this output was around 40 watts for me to drive the 813 rig to about 350 watts. My exciter put out the same power. What was going on?  

As an experiment, I took a 50 ohm 6 dB high-power attenuator (that had been wired-in under my operating position) and connected it between the new exciter and the 813 PA's RF input. When I keyed the rig, the PA's grid current rose to about 22 mA -- right around where it was when the Ranger was driving the rig. 

Hmmm... 

I poked around and discovered that the 6 dB attenuator I'd just tested with had originally been installed between the Ranger and the 813 transmitter. I'd forgotten about it, and I'd assumed that the Ranger had been directly driving the 813 rig with 40 watts, when in reality it had been driving the 813 transmitter rig with one-quarter of this power! Doh. Dope slap!  

An obvious solution was to keep the 6 dB high-power pad connected between my exciter and the 813 PA Deck's input, but this seemed like a waste of a good attenuator (high-power attenuators are expensive, after all). Was there another, simpler, way to decrease the output power of my exciter by a factor of 4? 

A Slight Change to the Design... 

The Exciter's voltage for the FET Drains was 26 volts. If I halved this value, then, in principle, I ought to get a quarter of the power (power changes with the square of voltage). 

Luckily, the design already has a 12V switching regulator (rated to 3 Amps), so I just moved the connection for the FET Drain power from 26V to the output of the 12V switching regulator. Keyed it up, and, voila, it worked! The meter readings for the 813 were where they were when driven with the Ranger. 

Length Matters... 

One interesting phenomena that I noticed when doing this, though: during my initial testing, I'd connected the Exciter's RF output to the PA's RF input through two lengths of RG-58 coax (because I'd originally placed the 6 dB pad between these two lengths) for a combined length of about 9 feet. Later, when I shortened the total coax between the Exciter and the PA Deck from about 9 feet to 3 feet, the PA Deck's Grid Current started reading in the 35 mA range rather than in the original 20 mA range and the Exciter's Drain current jumped from about 0.8 A to 1.2A. Neither of these were desired changes, so I went back to my original 9 feet of coax to interconnect the Exciter to the PA Deck. 

Why does a change of 6 feet make such a difference in operation? At the moment, I don't know. However, as the length of the interconnecting coax is shortened, Exciter Drain current increases (from about 0.8A with 9 feet of coax to about 1.2A with 3 feet of coax), so the implication is that, with shorter coax, the Exciter is seeing a lower load resistance. This then implies that the PA Deck's RF input doesn't look like 50 ohms resistive, and thus there is an impedance transformation taking place via the 50 ohm coax. 

(I connected an HP 3577A network analyzer to the exciter-end of the coax feeding the PA Deck. With the PA grid tuning set to peak grid current (when the exciter was connected), I made the following measurements:

• 9' coax: S11 mag: 0.92, S11 angle: -13.2
• 4.5' coax: S11 mag: 0.67, S11 angle: -24.1

When converted into a parallel representation of real and imaginary impedance components (because, after all, the Exciter's tank consists of parallel-connected components, not series), the resulting values are:

• 9' coax: Real: 47.4 Ω, Imaginary: -j202 Ω
• 4.5'coax: Real: 36.8 Ω, Imaginary: -j82 Ω

Assuming that the Exciter tank is tuned to compensate for the imaginary component, the Exciter tank sees a lower resistive component with shorter coax, which correlates with the increased Drain current that I see, and the resistive component with 9' of coax is quite close to 50 ohms. 

However, there is one puzzle that I don't yet understand: with 4.5' of coax, the imaginary component represents more parallel capacitance than that of the 9' coax, yet I find that, when tuning the Exciter's tank when using the 4.5' length of coax, I need to turn the Exciter's Tank capacitor to full-mesh (i.e. high-capacitance) for best-looking Exciter RF. Why do I need to add more exciter-tank capacitance when I've already added more capacitance at the exciter load? It doesn't make sense to me. Should I be working with the series-form of impedance instead (in which the impedance measured at the end of the 4.5' length of coax has less capacitance than the 9' length)? Have I made a mistake in my measurements? I don't know. 

Well, something to research on another day... 

Other notes: 

Note 1: If I'd kept the 6 dB attenuator connected between the Exciter and the PA Deck, then the effect of coax-length on Exciter performance would be less of an issue, because the attenuator would have "buffered" the effect of the PA Deck's input impedance on the Exciter. 

Note 2: There is interaction between the Exciter's Tuning capacitor and the PA Deck's Grid Tuning capacitor; the position of one will affect the other. That is, the amount of "junk" on the Exciter's RF waveform (monitored at the front-panel BNC) will change, depending upon how the Grid Tuning capacitor is changed. When tuning the transmitter:

1. First I peak the PA Deck's Grid current.
2. Then I adjust the Exciter Tank tuning for best looking RF at the Exciter's output (as observed at the Exciter's front-panel BNC). This is typically at, or near, minimum Drain current, as measured on the Exciter's front-panel meter.

Here's a screen-shot of bad-looking RF from the Exciter. Its tank needs tuning!

   

(Click on image to enlarge)

And here's the Exciter RF with its tank properly adjusted:

   

(Click on image to enlarge)

(Exciter RF waveform measured at the front-panel BNC, J6, using a Tektronix TDS320 scope (100 MHz bandwidth).) Schematics. There are four pages. Here they are:

   

(Page 1. Click on image to enlarge)

Notes on page 1: This page is essentially the same as the original design, but changes are:

• DC Voltage for IRF530s changed from 26 VDC to 12 VDC.
• 510 pf added to tank circuit (3570 pf total) so that the Tank circuit, when operating at 3.87 MHz, is properly tuned with Tuning Capacitor C11 at about half-mesh.

(Page 2. Click on image to enlarge)

Notes on page 2: 

No change from the original circuit. But because the LM2576 switching-regulator now must deliver an additional 800 mA (or so, to power the PA FETs), the inductor L3 really should be changed from 1000 uH to 470 uH or 330 uH. But it seems to run fine with the original value of 1000 uH, so I'll leave modifying this for another day. 

And, strictly speaking, I didn't need to incorporate a sequencer into the Exciter's design -- I could have used the existing sequencer in the 813 transmitter to perform the same function. But incorporating this sequencer allows me to easily test the Exciter as a stand-alone unit.

    

(Page 3. Click on image to enlarge)

Notes on page 3: 

This is the audio driver which drives the 813 Modulator Deck. Externally, and prior to this stage, I use a Behringer Xenyx 802 mixer/amplifier to amplify and equalize my microphone. 

For 100 percent modulation, the Modulator Deck requires an input level of about 80 volts RMS (when driven with a sine-wave -- this is about 226 Volts peak-to-peak). The simplest way to get this sort of amplitude is with a transformer. On eBay I found an audio output transformer (designed to present to a push-pull driving stage a load of 6.6K or 8K ohms when driving either a 4, 8, or 16 ohm load -- its Part Number is OT20PP), and I decided to connect it in reverse to drive my Modulator Deck so that I could transform the high-impedance of the Modulator Deck input to a low-impedance, and then drive this low-impedance with a speaker amplifier designed to drive loads in the 4 to 16 ohm range. 

To test which combination of input/output windings would work best in my application, I connected the transformer to the Modulator Deck and drove it with a stereo amplifier. With a 1 KHz sine-wave test signal, for full modulation (corresponding to an audio drive of about 80 Vrms into the Modulator deck), I needed about 12 Vpp of drive from the stereo amp. 

For the actual transformer driver, I used an LM1875 speaker amplifier. Its output is single-ended, so, to get 12 Vpp out with some headroom, I used the 26 VDC power supply to power it. 

I also decided to use the 16 ohm tap as the primary (driven by the LM1875) and the 6.6K ohm taps as the secondary (to drive the Modulator Deck). This is the lowest step-up turns-ratio provided by the windings, and the Modulator Deck's input impedance is transformed to be about 5.4 ohms, as measured at the output of the LM1875, which conveniently lies between the LM1875's 4 ohm and 8 ohm load specs. (Any other combination of windings would have resulted in a lower load impedance for the LM1875). 

When driving the Modulator Deck to full modulation, the LM1875 delivers about 3.6 watts into this 5.4 ohm load. 

There is a potentiometer to allow some amount of gain adjustment, but the primary gain is back at the Behringer mixer. And there's a mute circuit to mute the audio drive to the Modulator Deck when the 813 Transmitter is not transmitting. (The 813 transmitter does not like it when the modulator and modulation transformer are driven when the PA deck is not generating RF). 

The low-frequency -3 dB point is about 280 Hz (determined by R28 and C44), which I purposefully added when I discovered that lowering this frequency caused the AM signal to sound a bit fuzzy (due to IMD products related to the voice frequencies below this point). The upper -3 dB point for the exciter/813 rig (combined) is around 4 KHz. These points were measured by driving the modulator with sine-waves and measuring the peak-to-peak envelope of the modulated RF. 

As a precaution against EMI problems involving RF interacting with the audio components, the audio components are all placed within a separate shielded chamber (made using double-sided PCB stock) within the chassis. All signals which transition into this chamber from the area containing the Exciter's RF stages are first filtered using feed-thru caps and L/C (or R/C) low-pass filters.

    

(Page 4. Click on image to enlarge)

Notes on page 4:

1. The 26 VDC power supply is a Cosel 24V supply (adjustable +/- 10%), rated at 4.5 ADC that I picked up from eBay. Now that I've discovered that I don't need 40 watts of RF power, this supply could actually be rated at a much lower DC output current, but hey, hindsight is 20/20.
2. The AC Connector and AC Line filter are actually an integrated modular unit.
3. The VFO is an N3ZI DDS2 VFO. Its output is only about 380 mVpp, so I bump it up to about 2 Vpp (to drive the 'HC86 XOR gates) with an OPA690 op-amp. The 50 ohm resistor in series with the output was added to reduce some high-frequency ringing I had observed, but I'm not sure it's really needed -- I may have been mistaken in this measurement.
4. The VFO Amp is only turned-on when transmitting. With the chassis buttoned-up, I've found that, even though the DDS VFO is always active (even during receive), I cannot hear it on my receiver.
5. The Drain Current meter is 1.5 mA full-scale. The resistors (and sense-resistor) scale the current reading so that the meter represents actual current ÷ 2000.
6. For adjusting the Tank's tuning capacitor, I added an RF tap (R32 and R33) which connects to a BNC on the front panel. The series-2K ohms represented by R32 and R33 help to isolate the tank circuit from the capacitance of coax-cables used to connect this BNC to a scope.
7. And a diplexer is still used to help clean up the Exciter's RF output. The 50 ohm load for the Diplexer's parallel L-C circuit (i.e. the load for out-of-band frequencies) is actually seven 357 Ω, 1/4 watt resistors in parallel. And I placed the series L-C part of the diplexer in a Pomona box because I was concerned that, if not shielded, unwanted RF components would couple around it to the output.

813 Transmitter Wiring Diagram with the K6JCA Exciter installed.

(Click on image to enlarge)

And here are some photos! 

The Audio stage.  Note the shielded compartment. And the LM1875 amplifier attached to the side of the chassis for heatsinking.

(Click on image to enlarge)

In the rack and on the air!

   

(Click on image to enlarge)

(Note: This shot was taken with the FETs powered with 26V, rather than 12V, and a 6 dB attenuator between the Exciter and the PA Deck. With a 12V FET power source, the meter needle is about 0.4 mA (out of 1.5 mA FS), representing about 0.8 Amps of Drain current. 

Additional Notes: 

 Because the tank transformer is 1:1, I wondered what the effect would be if I moved the Tuning Capacitor (C11) from the primary side of the tank to the secondary side. This would allow me to more easily mount the cap, because it no longer would need to float. However, when I performed this experiment, I discovered two issues:

• The tuning range narrowed.
• Output power varied slightly with frequency.

Neither of these outcomes were positive, so I kept the tuning cap on the input side of the tank transformer, and I mounted it on a piece of polycarbonate plastic (from Tap Plastics) to isolate it from the chassis. 

 Resources: 

 Datasheets:

• LM1875 Datasheet
• OPA690 Datasheet
• OT20PP Transformer
• Cosel PAA100F-24 Power Supply

Caveats: 

1. I could have easily have made a mistake, so please regard (and use) this design accordingly. 

2. High voltages can kill. Use caution.

TX Overshoot on Flex 5K

[Additional Notes:

Latest News, 9 MARCH 2011:

Per FLEX, this problem is now FIXED!!!

Gerald's Email to FlexRadio Customers states, in part:

"Fixed ALC overshoot and corrected leveler gain
target in the transmitter audio signal chain.
These changes have been verified by customers
on air and in the FlexRadio lab using a digital
storage oscilloscope to eliminate overshoot."

7 MARCH 2011:

The word from Flex is that they've tested a solution which looks very promising.

Also, additional analysis reveals that the overshoot isn't due to Gibbs Phenomena (as I first hypothesized (see below)), but instead I now believe it's because ALC action occurs before the "real" TX signal is converted to I and Q. Shifting a signal's frequency components each by 90 degrees can result in a time-domain signal whose amplitude exceeds that of the original ALC-limited signal, as demonstrated in the graph below:

(Click on image to enlarge)

Blue Waveform (Sawtooth)
= sin(2πx) + 0.5(sin(2π2x)) + 0.33(sin(2π3x)) + 0.25(sin(2π4x)) + 0.2(sin(2π5x)) + 0.167(sin(2π6x))

Red Waveform (Sawtooth w/components shifted 90^o)
= cos(2πx) + 0.5(cos(2π2x)) + 0.33(cos(2π3x)) + 0.25(cos(2π4x)) + 0.2(cos(2π5x)) + 0.167(cos(2π6x))

[By the way, despite the way it looks, the Red waveform does not have a DC shift. If you examine the equation from which this graph was generated, you'll note that there are only cosine terms in the equation, and that there is no "constant" term, which would represent a DC shift.]

Imagine that the Blue waveform, being our TX signal, had been "normalized" by ALC operation to have a maximum amplitude of 1.0 (i.e. divide all the values in the graph above by 1.5 so that the Blue waveform has a Vpeak of 1.0). The Red waveform, which is the TX signal with its frequency components shifted by 90 degrees, would have a peak amplitude exceeding 1.0 if its components were also normalized by the same amount (divided by 1.5).

And so, to properly limit the signal, ALC must be applied after the conversion from real to complex data. Which, coincidentally, my fix (described below) does.

Interestingly, if a Triangle-wave's components are phase-shifted by 90 degrees, the resultant signal has an amplitude less than the original's amplitude. Which correlates nicely with our observation that we don't see overshoot with a Triangle wave as our signal, but we do if the signal is a Sawtooth wave.

Below is a plot from a MATLAB simulation.  It shows the following:

o  Audio sawtooth waveform, fundamental frequency of 200 Hz and band-limited between 200 and 2400 Hz, and of peak amplitude = 1.
o  Audio sinewave (200 Hz) of peak amplitude = 1 (will be used as reference).
o  The original audio sawtooth waveform, but now with its frequency components shifted by -90 degrees.

o  RF Output, LSB, with 200 Hz tone modulation overlayed (for comparison) on RF LSB waveform with sawtooth modulation (and both shifted +4 to that, when plotted, they won't cover our original audio waveforms).

o  RF Output, USB, with 200 Hz tone modulation overlayed (for comparison) on RF USB waveform with sawtooth modulation (and both shifted -4 to that, when plotted, they won't cover our original audio waveforms).

(Click on image to enlarge)

Note that the Sawtooth's RF waveform has a peak of 1.759 compared to the 200 Hz Tone's RF peak of 1.000, even though both the Sawtooth and the Tone have the same peak audio amplitude of 1.0.  Therefore, if a transmitter were adjusted so that, when driven with the tone, its output was 100 watts (peak), then if the transmitter were driven with a Sawtooth of equal amplitude to the tone, its peak RF output would be 310 watts (i.e. an increase of 4.9 dB in peak power).

And here is the MATLAB code for the simulation, for reference:




clear
clc

% k6jca SSB Modulation Example
%
% Let audio be a 200Hz Sawtooth, of maximum amplitude 1 and band-limited
% between 200 and 2400 Hz.

% Plot 10K points
for inc = 1:100000
    % time tick represents 0.1us.
    % So total simulation time will be 1e-7*1e5 = 10ms. (2 cycles of 200Hz)
    t = (inc-1)*0.0000001;
    x(inc) = t;

    % in-phase sawtooth waveform with frequency components
    % from 200 to 2400 Hz.  And scaled by 1.725 so that peak amplitude is 1.
    i(inc) = (sin(2*pi()*200*t)+ 0.5*(sin(2*2*pi()*200*t))...
         + 0.33*(sin(3*2*pi()*200*t)) + 0.25*(sin(4*2*pi()*200*t))...
         + 0.2*(sin(5*2*pi()*200*t)) + 0.167*(sin(6*2*pi()*200*t))...
         + 0.1429*(sin(7*2*pi()*200*t)) + 0.125*(sin(8*2*pi()*200*t))...
         + 0.1111*(sin(9*2*pi()*200*t)) + 0.1*(sin(10*2*pi()*200*t))...
         + 0.0909091*(sin(11*2*pi()*200*t))...
         + 0.083333*(sin(12*2*pi()*200*t)))/1.7275;
     
    % quadrature generated by shifting in-phase with a 90 degree delay
    % (i.e. subtract pi/2)
    q(inc) = (sin(2*pi()*200*t-pi()/2)+ 0.5*(sin(2*2*pi()*200*t-pi()/2))...
        + 0.33*(sin(3*2*pi()*200*t-pi()/2)) + 0.25*(sin(4*2*pi()*200*t-pi()/2))...
        + 0.2*(sin(5*2*pi()*200*t-pi()/2)) + 0.167*(sin(6*2*pi()*200*t-pi()/2))...
        + 0.1429*(sin(7*2*pi()*200*t-pi()/2)) + 0.125*(sin(8*2*pi()*200*t-pi()/2))...
        + 0.1111*(sin(9*2*pi()*200*t-pi()/2)) + 0.1*(sin(10*2*pi()*200*t-pi()/2))...
        + 0.0909091*(sin(11*2*pi()*200*t-pi()/2))...
        + 0.083333*(sin(12*2*pi()*200*t-pi()/2)))/1.7275;

     % For reference (and comparison) generate a 200 Hz tone of peak ampitude 1
     i_tone_200(inc) = sin(2*pi()*200*t);
     q_tone_200(inc) = sin(2*pi()*200*t - pi()/2);

     % Now let's generate our RF.  First, in-phase.  Frequency is 1 MHz.
     rf_i(inc) = cos(2*pi()*1e6*t);
     
     % quadrature by delaying rf by 90 degrees (subtract pi/2);
     rf_q(inc) = cos(2*pi()*1e6*t - pi()/2);
     
     % Now let's calculate out in-phase and quadrature channels...
     in_phase(inc) = i(inc) * rf_i(inc);
     quad(inc) = q(inc) * rf_q(inc);
     
     
     % ...and then add or subtract them to make LSB and USB
     lsb(inc) = in_phase(inc) + quad(inc);
     usb(inc) = in_phase(inc) - quad(inc);
     
     % For reference, let's also make LSB and USB using the 200 Hz tone,
     % so that we can see if its peak RF differs from the sawtooth's 
     % peak RF
     lsb_tone(inc) = i_tone_200(inc) * rf_i(inc) + q_tone_200(inc) * rf_q(inc);
     usb_tone(inc) = i_tone_200(inc) * rf_i(inc) - q_tone_200(inc) * rf_q(inc);
     
end

top = max(i);
top_q = max(abs(q));

top_lsb = max(abs(lsb));
top_tone = max(abs(lsb_tone));

figure
plot(x,i,x,q,x,i_tone_200,x,lsb_tone+4,x,lsb+4,x,usb_tone-4,x,usb-4);
grid on;
legend('sawtooth (input)','input shifted -90 degrees','200Hz Tone (Reference)','LSB of 200Hz Tone','LSB of sawtooth','USB of 200Hz Tone', 'USB of sawtooth');




My original post is below...]

On the topic of TX overshoot with the Flex 5000...

Last summer a friend of mine mentioned that he was having a problem with PowerSDR and his Alpha amplifier -- it looked to him as though RF overshoots at the Alpha's input were causing it to shut down, and these shutdowns were occurring frequently enough to really annoy him.

I decided to do a bit of investigation. When I looked at the RF envelope from my Flex 5000, I saw overshoots with my 1.18 Console. Thinking that the problem might be related to the software revision, I downloaded the 2.0.7 PowerSDR Console and ran my tests again. Below are snapshots of the OUTPUT RF waveform from my 5000 using the 2.0.7 console.

Oscilloscope is my Tek 2445 that I use to monitor my transmit RF (via an RF-sampler). The two horizontal cursors you see on its CRT mark the peak-to-peak level of the RF signal when the 5000 is in TUN mode and the Tune level is set to 10 watts.

For the following snapshots, PowerSDR is set to:

• Freq: 3.863 MHz
• Mode: LSB
• Drive: 10
• Leveler: Disabled
• TX EQ: OFF
• DX: OFF
• CPDR: OFF
• DEXP: OFF

(Why use 10 watts and not 100? At some point the PA will limit and it won't be able to deliver any additional power. If we run the experiments at high power, we risk having PA-limiting influence our results. But if we run our experiments at low power, we should be able to get a better representation of how high the peaks actually go, because they won't be subjected to PA limiting.)

Anyone can try this experiment with these setting. Here are my results:

First, two nicely behaving waveforms:


Notice how the peaks don't exceed (by much) 10 watts?

But look at these next two. Yikes!



The four snapshots are:

1. Triangle waveform from internal internal software Transmit generator. Looks great!
2. Pulse, from the internal software Transmit generator (at its default settings for Pulse mode). Looks great!
3. Sawtooth waveform from the internal software Transmit generator. Yikes -- it greatly exceeds the 10-watt cursors! Peak power is 49 watts on my LP-100 power meter, yet DRIVE is set to 10 watts.
4. My voice, saying "Ahhh" loudly into microphone. Yikes again! Peak power is about 40-50 watts on my LP-100 power meter, yet DRIVE is set to 10 watts.

Both the "Ahhh" and Sawtooth signals greatly exceed the 10 watt Drive setting on the 2.0.7 console, and the level of the voice signal is very similar to what I'm seeing with my modified 1.18.4 console.

In AM mode, the sawtooth (as modulation) looks exactly as it should, so I don't believe the issue with the sawtooth is that it's somehow "broken."

I thought I'd experiment with the code a bit and see if I could gain further insight...

First, I experimented with the ALC code itself, as my first suspicion was that the overshoot was due to the non-zero attack time. But nope, that wasn't it. I changed the attack time to 0 (so that the ALC was a true peak-detector, and not one with a non-zero attack time), and it made no appreciable difference in the gross overshoots experienced by voice or sawtooth SSB modulation.

I then played around with the location of the ALC algorithm, and the results are very interesting. If the ALC code is prior to the filter_OvSv routine (as it is in the version of code I'm using (v1.18.4) as well as in the SVN 3862 code that I downloaded), then both Voice and Sawtooth waveforms grossly overshoot the target power, per the photos above. In other words, the current ALC code in its current position, in either my console (1.18.4) or in the 2.0.7 console, does not properly limit the output RF for certain waveforms.

However, if I move the ALC code to just after the filter_OvSv routine (and change the buffers used for ALC in newDttSPAGC from buf.i to buf.o), then there is minimal overshoot for all waveforms (Tone, Triangle, Sawtooth, and Voice), and their peaks are all near the target power. In other words, the output RF looks beautiful. [Note: I wasn't able to test the PULSE waveform, because my 1.18 version of the console doesn't have PULSE as one of the options for the test generator].

I'll make a wild guess that the overshoot is due to the bandpass filtering in filter_OvSv and thus similar to Gibb's phenomena. But this is just a guess. Conceptually, it makes sense to me that the ALC processing should be as late in the processing chain as possible, and moving it to just after filter_OvSv seems to be a better place for it than prior filter_OvSv. However, I'll be the first to admit that I don't know what the ramifications are of doing this. Are other problems introduced? I don't know.

Building an 80-Meter Class E/F RF Amplifier...

After simulating on my computer an 80-Meter Class E/F amplifier (here and here), I decided to actually build one. Here it is:

(Click on image to enlarge)

With a 26V power supply, RF Power Out is about 40 watts. And efficiency of the MOSFET final is in the range of 85-95% (depending upon which power meter I use to measure RF power).

In the image above:

• The two MOSFETs (IRF530, N-Channel) are mounted on the vertical copper plate at the upper-left.
• The Tank circuit, including the transformer (5-turn air-wound coil) and a variable air capacitor (for tuning the tank) are at the upper-middle.
• Control logic (and Symmetry adjustment potentiometers) are at the middle-left.
• Two solenoid-style inductors (MOSFET Drain DC feeds) are lower middle.
• A 12V switching voltage-regulator is at the lower-right (with the large toroidal inductor).
• You can also see the two scope-probes attached to the MOSFET Drain busses, and the RF output is the BNC at the right.

Here are the schematics:

(Click on image to enlarge)

(Click on image to enlarge)

Notes on the Design:

Page 1:

This page contains the MOSFET Amplifier and its driving circuitry.

The amplifier consists of a pair of IRF530 MOSFETs in a push-pull configuration that drive a tank circuit consisting of transformer T1 and parallel capacitors C7 and C10. Why IRF530 MOSFETs? They were in my junk box!

These MOSFETs are rated at 100 volts max V_DSS and have an R_DS(on) of about 0.18 Ω (the latter depends upon which manufacturer's datasheet you look at).

The tank transformer, T1, consists of two windings, 5 turns each, with the secondary winding wound inside the primary winding (the windings are concentric). Total coil length is about 2.75", and the inner-diameter of the outer coil is about 1.5". After I wound T1 I discovered that its inductance measured to be around 570 nH. I'd been shooting for about 400 nH, but the difference isn't a big deal -- 570 nH just lowers the overall Q a bit (from a simulated Q (with 400 nH) of 5 to an actual Q of 3.6 at 3.87 MHz with a 50 ohm load). (See note later in this posting regarding measuring inductance of an unknown inductor).

C7, which, in combination with C10, forms the resonant tank capacitance, is actually six 510 pf ceramic capacitors (low ESR caps from American Technical Ceramics (their 700B series)). C10 is an air variable (20-420 pf) from my junk box, and it gives me a tuning range of approximately 3.78 MHz to 4.03 MHz.

Peak voltage across these tank components is on the order of 80 volts or so, so there's no reason for high-voltage parts. However, current through the inductor and capacitors is on the order of 3 to 5 amps RMS (per my SPICE simulations), so low-ESR components are highly recommended.

The MOSFET Drains are fed via 18 uH inductors L1 and L2. I had wanted to use higher inductance, but I didn't have anything in the junk box that was suitable (high inductance and high self-resonant frequency (S.R.F.)). However, I did have a couple of J.W. Miller 5252 inductors -- these are 125 uH inductors, but their self-resonant frequency is only about 2.5 MHz. I removed the top two winding layers (of three total layers) to give me an inductance of 18 uH and an S.R.F. of about 48 MHz.

I decided that the simplest way to drive the PA MOSFETs would be with MOSFET drivers. I chose IXDD414 MOSFET drivers (note: these are obsolete parts, but I found mine on Ebay). They drive the MOSFETs via 1-ohm resistors, which seemed to reduce ringing (but this observation really should be reconfirmed -- take it with a grain of salt).

To drive the IXDD414 Drivers I use two XOR gates to generate, from the VFO signal, two signals of the same frequency but 180 degrees out of phase with each other. One of the XOR gates inverts the VFO signal, while the other passes it through uninverted. Both XOR gates are on the same die, which should minimize the differential delay between the two signals.

The VFO signal is AC-coupled to an input on each of these two XOR gates, and two potentiometers provide variable DC offsets to each of these same two inputs so that the duty-cycle (i.e. symmetry) of each XOR output can be independently adjusted.

(Note: For testing I drove the XOR gates with an HP 8640B signal generator, set to +10 dBm. At this level, this generator provides a nice, very low distortion sine-wave with about a 4 Vpp amplitude (rather than 2 Vpp, which you'd expect at +10 dBm, because the 8640B is now terminated in a high impedance, rather than 50 ohms)).

The final bit of circuit on this page of the schematic, Q1 (a P-channel MOSFET), ramps the DC voltage feeding the IRF520 Drains up and down when the amplifier is turned on and off, thus providing a soft, rather than hard, transition to the output RF envelope. (I tried using the enable pins on the IXDD414 ICs in lieu of adding this MOSFET, but I found there were voltage spikes on the Drains of the IRF530 MOSFETs exceeding their V_DSS when I transitioned these Drivers ON using their enable pins).


Page 2:

This page contains the voltage regulators and control circuitry.

An LM2576 switching regulator provides the +12VDC power for the IXD414 MOSFET Drivers. These two drivers require about 0.4 - 0.5 amps of current, total. A 12V linear regulator would have had to dissipate about 5 to 6 watts of power, which is why I chose to go with a more efficient switching regulator.

The values for the switching-regulator's components are straight out of the LM2576 datasheet. Note that this datasheet specifies a 1000uH inductor for the 12V regulator when the input voltage is around 26V and the load current around 0.4 Amps. Pulse Electronics has a series of "50 KHz Inductors" (available through Digikey) that are recommended for LM2576 applications. I used the PE-53120, which is a 1000 uH inductor.

A 7805 5-volt linear regulator provides 5VDC for the digital logic.

This amplifier is designed to be keyed by my 813 AM transmitter. Because I want the VFO to be inaudible in my receiver when I'm not transmitting, I need some way to disable it (or move it off frequency) when I'm not transmitting. Also, I want the VFO to be enabled and generating its signal before I apply power to the MOSFET Drains and to go OFF after I remove power from the MOSFET Drains when I'm done transmitting, so that the VFO is stable at all times while power is being applied to the MOSFETs. In other words, I want to "nest" the MOSFETs' ON/OFF cycle within the VFO's enable/disable cycle.

Although I could use the 813 Transmitter's sequencer to nest the MOSFET Power ON/OFF within the VFO Enable/Disable, for ease-of-testing I decided to incorporate a sequencer into this design to allow me to test this amplifier as a stand-alone unit. This new sequencer is also based upon the W2DRZ design. In the future, when I incorporate this circuit into the 813 AM Transmitter, I will decide if I should use the 813 AM Transmitter's sequencer in lieu of this one.

This sequencer runs at a 50 Hz clock rate because I want it to run faster than the sequencer in my 813 Transmitter (which I've set to run at roughly a 10 Hz clock rate).

I plan to drive this amplifier with an N3ZI DDS Module, which I would like to disable when I'm not transmitting so that it doesn't interfere with reception. There are a couple of ways that this might be done. One is to shift the VFO to a completely different frequency (using the DDS Module's "VFO B/A" input).

Moving the VFO off frequency when not transmitting would prevent Receive interference, but it might be also useful to shut off the VFO when transmitting so that the MOSFET Drivers (which are driven by the VFO via the XOR gates) aren't consuming power from the 12V supply. There are several ways in which one might do this:

One way might be to "lift" (via an open-collector/drain driver) the ground-end of R1, the 6.8K resistor attached to the AD9834's FS ADJUST pin (pin 1), when not transmitting. Per the AD9834 datasheet:

IOUT_{FULL SCALE} = 18 * FSADJUST/R_SET
(FSADJUST = 1.15V nominal, and
R_SET is R1 in the N3ZI DDS2 VFO schematic)

Thus, raising the resistance of R1 during Receive should reduce IOUT to near zero.

Another possible way to disable the VFO might be to simply short the IOUT output from the AD9834's IOUT to ground. This is a current-source output, of which the DDS chip has two (IOUTB is the second one), and the datasheet states that IOUTB can be shorted to ground if not in use, so I would think that one could also short-out the first output, too, to disable the DDS (both are rated at the same output current). But is this advisable? I can't say. Also, the current at IOUT would still be generated and thus sourced out the IOUT pin, and so any "non-zero area" current-loop formed by the "shorting" components and IOUT's signal path could still create some amount of interfering RF emissions on the receive frequency.

And perhaps the easiest way would be to simply take the IXDD414s' EN pins (pin 5) to ground at the same time that the VFO frequency is shifted (i.e. VFO disabled).

There are some signals on this schematic page, such as MUTE VFO, that could be used for just this purpose (e.g. tie MUTE VFO to the U8.5 / U9.5 node). I'll determine which route is best when I incorporate the N3ZI DDS module.


Notes on Construction:

I built the circuit on a piece of 6.5" x 4.5" scrap single-sided copper-clad FR4 circuit board. The copper plane on this board is used as the circuit ground. I cut up pieces of double-sided FR4 PCB material to use as mounting pads and power busses (one side of each is soldered to the copper "ground plane" on the main board).

Regarding the MOSFETs and their MOSFET drivers (and any other components with high slew-rate signals), it's important to minimize parasitic inductance, so keep leads as short as possible.

Caps used for power-supply bypassing (e.g. C23-C26, C29 and C30) should have very high self-resonant frequencies, and they should be mounted as close to the IXDD414s power pins (and ground) as possible to minimize unwanted inductance from their leads.

It's a good idea to try to ensure that each Drain of the IRF530 MOSFETs sees approximately the same amount of capacitance-to-ground from the wiring and other sources -- my SPICE simulations showed that unequal amounts of capacitance-to-ground for the two MOSFETs results in unequal voltage peaks, with respect to ground, at the Drains of the MOSFETs.

The IRF530 MOSFETs are attached to a heatsink consisting of a copper buss-bar that's 1" x 4.75" x 0.125". This heatsink is electrically attached to the circuit ground with copper tape and also a small angle bracket that mechanically holds the heatsink to the circuit board.

The transformer T1 is constructed of rectangular 3/16" x 1/16" enameled copper magnet wire (this is the same wire used by Taniguchi, Potter, Rutledge in their 200 W Power Amplifier which appeared in the Jan/Feb 2004 issue of QEX). It consists of two windings, 5 turns each. Total length is about 2.75", and the inner-diameter of the outer coil is about 1.5". The secondary-winding is closely wound inside the primary winding (to minimize leakage-inductance), and I covered one of the windings with Kapton tape when I found that the wire's enamel was sometimes cracking as I bent it, and I was concerned that the two windings might short-out to each other.

The LM2576 datasheet has useful tips for layout and interconnection of components.


Adjustments and Tuning...

There are three adjustments to make: The two Symmetry potentiometers, R3 and R6, set the duty-cycle of the outputs of the two XOR gates, and the Tune variable capacitor, C10, in the output Tank Circuit allows the Tank's resonant frequency to be tuned roughly 250 KHz.

Initial Setup:

• While not transmitting (PTT IN is still high), I first set the two pots, R3 and R6, to midway in their adjustment range. While monitoring the XOR outputs with a scope, I then I tweak the two potentiometers so that the outputs of the XOR gates are each at (or close to) a 50% duty-cycle.
  

• I then take PTT IN low to enable the transmitter. While monitoring the two Drain waveforms on an oscilloscope, I tweak the two pots to try to minimize the "ringing" on the MOSFET Drain signals. (See note below regarding how to accurately probe the Drains). Note: The final setting of each pot seems to correspond to a duty-cycle very close to 50% for each XOR output.

• Then I adjust the frequency of the VFO for minimum Drain current (using a voltmeter across the 0.1 Ω resistor, R12, to monitor DC Drain current).

• I again tweak the pot adjustments to again minimize ringing on the two Drain waveforms. (If you have a Spectrum Analyzer, you can also monitor the spectrum (out to, say, 100 MHz) and adjust the pots to minimize the higher-frequency spikes).

The result should look something like this:

(Click on image to enlarge)

Once these adjustments have been made, I find that I can keep the pot settings fixed and only change C10 when I move to a different frequency. (When moving to a different frequency, you can try retuning C10 for minimum Drain current, but I personally find that approach only gets me into the ballpark. The approach I follow is described in more detail just a bit later in this section.

Per my measurements, the efficiency of the MOSFET amplifier itself is around 90% (this number should be taken with a grain of salt, as it depends a great deal upon the accuracy of your watt-meter as well as DC current and voltage measurements!), and efficiency drops to about 86% at about 100 KHz of either side of the frequency to which C10 is tuned to. However, ringing is starting to look pretty severe at this point, and so I'd recommend changing the VFO by no more than, say, +/- 50 KHz before retuning C10.

When misadjusted, the Drain waveforms can look like the following image (or much worse!). Note that this is ringing on the waveform, not oscillation.

(Click on image to enlarge)

When changing frequency, I only adjust the capacitor, C10. The two pots I leave alone after their initial adjustment (see above).

When tuning to a new frequency, my procedure to adjust C10 is a bit iterative. You can try to adjust it for minimum Drain current, but I find that this approach doesn't always work. Instead, I do the following:

• Rotate C10 to where you think it should be, approximately, for the frequency you'll be using.

• Tune the VFO for minimum Drain current.

• If the frequency of the VFO, after tuning, isn't close enough to where you'd like to be, tweak C10 a bit and repeat.

This approach seems to work for me.


Knocking Down the High-Frequency Junk

The output of the Class E/F amplifier is fairly dirty with high-frequency trash from 1) Harmonics (and ringing) on the MOSFET Drain waveforms, and 2) pickup of harmonics from the XOR gates and MOSFET Drivers.

Here's the spectrum of the Amplifier's output prior to adding an external filter.:

(Click on image to enlarge)

(Measurements made through a 30 dB Bird attenuator. The signal at the far left is the fundamental (at about 3.8 MHz), and the next signal (very low level) is the second harmonic.)

The above chart shows spurs out to 100 MHz. In reality they extend well past this frequency. In an attempt to reduce them, I first tried a 5-element Chebychev low-pass filter from the tables in the ARRL Handbook (Fig. 12-19, #7, for f_co of 4.5 MHz, which results in values of 620 pf for the input and output caps, 1200 pf for the middle cap, and 2.39 uH for the two inductors).

But when I tried using this filter, I had to crank up my power-supply to about 30 volts to get the same amount of RF power output. So I nixed this filter (The peak Drain-Source voltage for the IRF530 MOSFETs was at its limit).

Instead, I decided to incorporate a simple diplexer, similar (in topology) to the one described in the Part 2 article of the High-Efficiency Class-E Power Amplifiers. . To keep the design simple, I designed for a Q of 1, which means that the inductors and the capacitors are the same in both "halves" of the diplexer. Here's its schematic:

(Click on image to enlarge)

The caps are dipped silver-mica (300V rating), and the inductors were wound with 26 gauge enameled wire. I'm using a BNC-mounted Tek 50 ohm, 1/2 watt termination to terminate the parallel L-C branch of the diplexer. After about 1/2 hour of "key-down" operation, this termination gets a little bit warm.

With this filter, the PA requires a DC Power Source of about 25.7 VDC to generate 39 watts RF Power out.

And here's the resulting spectrum:

(Click on image to enlarge)

Because of its low Q, the Diplexer doesn't do much to knock down harmonics that are near the fundamental, such as the 2nd and 3rd harmonics. But because of the symmetrical output of a push/pull amplifier topology, the 2nd harmonic is already quite far down (when the Symmetry pots are properly adjusted), and the 3rd harmonic is now more than 40 dB below the fundamental.

One advantage to a diplexer implementation is that it presents a near 50 ohm load to all frequencies (assuming that the external load presented to the series L-C filter is 50 ohms at its resonant frequency). This can help stabilize an amplifier that might otherwise be unstable when presented with off-frequency unknown impedances.


Accurately Probing the MOSFET Drain Voltage Waveforms

If you try measuring Drain voltage waveforms with a standard scope probe and its ground lead, you are not going to get an accurate picture of what the waveform actually looks like. Instead, you are very likely to see all kinds of junk on the signal. This "junk" really isn't on the signal -- it's radiated noise picked up by the inductive "loop" formed by the scope probe and its ground lead.

To get an accurate picture of how the signal really looks, you need to minimize the capture area of this loop. The best way to do this is to keep the ground "lead" as short as possible, and connect ground to the probe as close to the probe tip (where the signal being probed is) as possible.

For this purpose I adapted a Tektronix PCB to probe-tip adapter (Tek p/n 131-4244-00, made for probes such as the P6139A). I soldered two pins of the four-pin "ground" shell to the PCB copper-clad upon which I'd built the amplifier (the other two pins are in the air, as you can see in the photo below), and the socket for the probe tip I soldered to the Drain buss for one of the MOSFETs. The other MOSFET has the same setup so that I can monitor both Drains simultaneously.

(Click on image to enlarge)

(There's a useful app note from Analog Devices here. Although it discusses probing high-speed signals, its techniques are quite applicable to this circuit, too.)


Other Notes and Thoughts:

1. Measuring Inductance or Self-Resonant Frequency of an inductor:

Here's the technique I use. There might be better approaches, but this one worked for me...

(Click on image to enlarge)
2. Tank Capacitors:

Per my SPICE simulations, the tank circuit has large currents -- on the order of 3 to 5 Amps RMS. So you want the tank capacitance to have a very low ESR to minimize unwanted power dissipation.

For my tank capacitance, I used six 510 pf caps in parallel. The capacitors are manufactured by American Technical Ceramics, and are from their 700B series. The ESR of their 510 pf capacitor is in the range of 0.04 - 0.05 Ω. If we assume 0.5A of current passes through each cap (ball-parking 3A RMS total current through six caps), the power dissipation in each is then on the order of 0.01 watts. Not too bad!

(In hindsight, though, I probably should have used their 470 pf caps. The 510 pf caps have a working voltage of 100 volts, which is a bit too close to the voltage they're actually seeing in this circuit (see scope images above of the Drain waveforms). The 470 pf caps, on the other hand, have a working voltage of 200 V, which gives a nice margin. If I can get these caps in, I'll replace the 510 pf caps.)

3. IRF530 MOSFETs: These have a max V_DSS of 100V. I'd like to have a bit more margin (per the discussion above regarding the 510 pF capacitors), but that's what was in the junk box when I looked for suitable parts. We'll see if there's any issue with reliability.

4. Some useful formulas:

• For a resonant circuit: L = 1 / [ (2*π*f)² * C ], or C = 1 / [ (2*π*f)² * L ]
• For a series RLC circuit, Q = (1/R) * √L/C, which, when combined with either of the two equations above, gives us: Q = |X/R|.
• For a parallel RLC circuit, Q = (R) * √C/L, which, when combined with either of the two equations above, gives us: Q = |R/X|.
• Note that when Q = 1, R = |X| for both the series and parallel RLC circuits.
  

5. 24V vs. 12V Power Supplies:

This design could be run from 12V (e.g. 13.4 VDC) rather than 24V (actually 26V) by changing the design of T1 to transform the 50 ohm load to a lower resistance across MOSFET Drains. This approach has two advantages: 1) Lowers the peak-voltage across the Tank components and at the Drains of the MOSFETs, and 2) Eliminates the need for the 12V switching regulator.

However, with the 24V supply, I can adjust the supply's output voltage (typically +/- 10%) to vary RF Output Power independently of the 12V supply (in case there are other radios running on the same 12V supply that might depend upon that voltage being fixed). Also, use of a 24V supply means that the transformer T1 for this application has a turns-ratio of 1:1, which (in my mind) simplifies its design. It would need to be on the order of 1:4 for a 12V application.

6. Transformer T1:

T1 consists of two concentric solenoid coils, air-wound with thick wire to minimize losses. It's possible that this transformer could be wound on toroidal or balun forms instead, using smaller gauge wire than the very large rectangular magnet wire that I used. It would be interesting to see what the effect of such a transformer would be on overall efficiency. Perhaps someday...

7. RF Power and Efficiency versus Power Supply DCV:

(Click on image to enlarge)

Notes:

• DC Voltage measured at the Q1 side of L1 and L2.
• The burbles in the efficiency curve are probably due to measurement errors on my part.

References:

Datasheets:

• IXDD414 MOSFET Driver
  
• 74HC86 Quad XOR Gate
  
• IRF530 N-Channel MOSFET
  
• IRF9530 P-Channel MOSFET
  
• IRF9540 P-Channel MOSFET
  
• ATC 700B Series Capacitors
  

Misc:

• W2DRZ Sequencer
• N3ZI DDS2 Module
• Analog Devices App Note: High-Speed Time-Domain Measurements -- Practical Tips for Improvement
• Toroid Winding Site
  

Spice:

• Free Download of LTSpice here.
• LTSpice "Getting Started Guide" here.

Class E Amplifiers:

• Sokal, "Class-E RF Power Amplifier", Jan/Feb 2001, QEX
• Sokal, "Class-E High-Efficiency RF/Microwave Amplifiers", (a more detailed paper)
  
• Lau, Chiu, Qin, Davis, Potter, Rutledge, "High-Efficiency Class-E Power Amplifiers" Part 1, May '97, QST, and Part 2, June '97 QST
  
• Davis, Rutledge,"A Low-Cost Class-E Amplifier with Sine-Wave Drive"
• Der-Stepanians, Rutledge, "10-MHz Class-E Power Amplifiers"
  
• Melia, Robert, O'Conner, Class-E Power Amplifier Design
• Tayloe, Class E Amplifiers (Norcal QRP Presentation)
• Class E Design Software from Tonne Software (I've never used this, so use at your own risk)
• WA0ITP Class-E Amplifier Design Spreadsheet (I've never used this, so use at your own risk)
• Class-E AM Forum
• Class-E AM Transmitters (WA1QIX)
• Class-E Amplifier Experiments , Calculations, and Notes on Designing Class-E RF Amplifiers, all by Bill Slade. (I've not verified the accuracy of these posts.)
  

Class E/F Amplifiers:

• Taniguchi, Potter, Rutledge, "A 200 W Power Amplifier", Jan/Feb 2004, QEX
• Letters, May/June 2004, QEX
• Letters, July/August 2004, QEX
• Kee, Aoki, Hajimiri, Rutledge, "The Class E/F Family of ZVS Amplifiers"
• Kee, "The Class E/F family of Harmonic-Tuned Switching Power Amplifiers" (Cal Tech Thesis)
• Kee, Aoki, Rutledge, "7.1 MHz, 1.1 KW Demonstration of the New E/F2,odd Switching Amplifier Class"
• Jeon, "Design and Stability Analysis Techniques for Switching Mode Non-Linear Circuits: Power Amplifiers and Oscillators" (Cal Tech Thesis)
  
• Kee, et al. U.S. Patent, No. 6,724,255
• Niknejad, "Class E/F Amplifiers" (presentation, EECS 242, U.C. Berkeley)
• Bohn, Kee, Hajimiri "Demonstration of a Switchless Class E/Fodd Dual-Band Power Amplifier (a 40 and 30 meter dual-band amp)
  

Caveats:

1. I could have easily have made a mistake, so please regard (and use) this design accordingly.