Improving the Frequency Stability of the Icom IC-725 Transceiver

Summary:

My "stock" IC-725 transceiver's frequency drifts over time (significantly if one is operating any of the newer digital modes).  To improve stability, I made the following modifications to my IC-725:

1. Change the 9 MHz BFO from the stock crystal oscillator instead to a BFO generated with an  Si5351A Clock Generator IC on a pre-fabricated development board.  (The Si5351A, using one of its outputs, generates the four different BFO frequencies required for different modes of operation.).  The Si5351A is clocked with a Temperature-Compensated Crystal Oscillator (TCXO) in lieu of its normal 25 MHz crystal reference.
2. Change the 30.72 MHz Reference Oscillator from the stock crystal oscillator to an Icom CR-64 "High-Stability Crystal Unit", and power its heater via regulated +12 VDC (instead of the radio's 13.8 VDC supply) by adding a LM2940-12 low-dropout regulator to the transceiver's "PLL Unit" PCB.
3. Change the fan from only being on during Transmit to always being on anytime the radio is ON (to eliminate internal-to-the-radio temperature excursions over a relatively short period of time that might affect oscillator frequencies).

I wanted to have a dedicated WSPR radio that could be controlled by a Raspberry Pi computer.  Searching the shack, I dug up an Icom IC-725 radio that I had purchased new back in the late 80's (or early 90's), when I was getting back into ham radio.

It has a computer-control interface that I could drive with a USB adapter connected to the Raspberry Pi, and so it seemed to be the ideal rig.  And it hadn't been used for years, so why not put it back to work?

But when I attempted to make WSPR contacts with it, my reported drift-rate (as reported by stations copying my transmitted WSPR signal) was 4 Hz/minute!  Ouch!

Clearly, my stock IC-725 was not up to the task.  Which got me thinking...could I improve it?

IC-725 Oscillators:

There are two oscillators in the IC-725.  One is a 30.72 MHz oscillator used for frequency conversion (on the "PLL Unit" PCB), and the other is the BFO oscillator that operates around 9.01 MHz (actual frequency depends upon the mode of operation).

Both oscillators use crystals, but the BFO changes its frequency by switching various inductors and capacitors in series with the crystal to pull its frequency one way or another, depending upon mode.

Measurements I made showed both oscillators drifted appreciably over time and both oscillators would need improving.

Improving Frequency Stability of the 9.01 MHz BFO:

The BFO switches between the following frequencies, depending upon mode:

• USB: 9.0130 MHz
• LSB: 9.0100 MHz
• CW(Rx): 9.0098 MHz
• CW(Tx): 9.0106 MHz
• AM/FM: BFO Off

An obvious choice was to eliminate Icom's method of switching inductors & capacitors to vary the oscillator frequency and instead to replace it with a more modern part, the Silicon Labs Si5351 Clock Generator.

To make my task easier, I chose Adafruit's pre-assembled breakout board with an installed Si5351:

BFO Generation using the Si5351, Hardware:

BFO hardware changes consist of modifications made to the radio circuitry plus the addition of a separate Arduino/Si5351 clock-generator daughter-board.

First, the daughter-board description...

Clock-generator Daughter-board:

I wired a small daughter-board (to mount behind the radio's front panel) with an Arduino NANO processor and the Si5351 clock generator circuitry.  The Arduino monitors the IC-725 "mode" signals and, depending upon their state, programs the Si5351A to generate the appropriate BFO frequency.

The Si5351A came pre-assembled on a small PCB.  I replaced its "stock" 25 MHz crystal a 25 MHz TCXO to improve frequency stability over temperature.  (Note that the Si5351A is not spec'd for use with an oscillator module such as the TCXO I am using.  More on this later in the post.)

Schematic Notes:

1. Note that the crystal at location Y1 must be removed from the Adafruit Si5351 board.  U4, a 25 MHz TCXO, now provides the frequency reference for the Si5351.
2. The control signals to the Nano from the IC-725 (USB, LSB, CW, and RX) have logic-high levels equal to 8 volts.  Therefore, voltage dividers (each consisting of a 3K and a 4.7K ohm resistor) to bring the levels into spec with the Nano's input requirements.
3. Per the CTS datasheet, pin 6 of the 25 MHz TCXO (U4) should be filtered with a 100 nF capacitor.  However, note that the part I am using is not one whose frequency can be "voltage-controlled".  It's not clear if this capacitor is still required, but I installed it anyway, just in case.
4. Resistors R13 and R14 form a simple voltage divider to bring down the output level of the TCXO prior to its connection to the XA pin of the Si5351.  C12 rolls off the higher frequencies of the TCXO's square wave, and C13 capacitively-couples the signal to the XA input.
5. Capacitors C6-C9 provide one-pole EMI filters in conjunction with the input voltage-divider resistors, to reduce any RF that might be coupled onto these wires.
6. If U4's 25 MHz frequency shifts with aging of the oscillator, the Arduino software can program the Si5351 to offset this aging.

IC-725 BFO Circit Modifications:

Notes on the BFO Modifications:

1.  I removed crystal X1 and installed an SMA connector on the metal shield surrounding the BFO oscillator circuitry.

2.  A 10 nF cap connects the SMA connector to the base of Q31, which should now be acting as an emitter follower rather than an oscillator.  An output from the Si5351 is programmed to generate the appropriate BFO frequency and it drives Q31 via the SMA.

3.  The BFO frequency depends upon the state of the USB, LSB, CW, and RX signals, as follows:

 

       Mode            Frequency (MHz)

       LSB                9.0100

       USB                9.0130

     CW(RX)           9.0098

     CW(TX)           9.0106

     AM,FM             No Output

These signals are tapped off the IC-725 at the following locations on its "Main Board":

      Signal     Location

       LSB        U15 pin 4

       USB       U15 pin 2

         CW       U15 pin 3

         RX        D61, Cathode

(Note that if LSB, USB, and CW are all low, then the IC-725 is in AM or FM mode and the Si5351A is programmed by the Arduino to turn OFF the BFO.

Si5351 9 MHz BFO Spurs:

There are spurs on the 9 MHz signals, but, assuming the Si5351's output divider is set to an even number, the larger of these spurs lie outside the passband of the 9 MHz filters (i.e. they would mix with filter stop-band noise, rather than with within-the-filter's-passband signals).  Thus, they shouldn't be a problem.

(See the Arduino Code section below for more detail on these spurs and how I minimized (but did not eliminate) them).

Improving Frequency Stability of the 30.72 MHz Oscillator:

I tried a number of different ways to improve stability of the 30.72 MHz oscillator.  These included:

1. Using the Si5351A to generate the 30.72 MHz clock (abandoned due to, in my opinion, unacceptable spur generation).

2.  Using a Voltage-controlled Temperature-Compensated Crystal Oscillator (VCTCXO).  (Abandoned to due unacceptable frequency drift over time).

3.  Using the Icom CR-64 High-Stability Crystal Unit (performance deemed acceptable after modifying the IC-725 to power this unit with regulated +12 VDC in lieu of the "unregulated" 13.8 VDC power to the radio).  

(Note, the CR-64 is not an OCXO (Oven-Controlled Crystal Oscillator).  It actually is just a crystal packaged with an oven that replaces the original crystal of the 30.72 MHz oscillator -- the rest of the oscillator's circuit is still the original circuit of the radio, outside of the CR-64, and thus C134 and L35 still adjust the oscillator's frequency.)

I'll discuss the issues with the first two methods listed above later in this post.  Let's look at the method I finally chose:

IC-725 Modifications to the 30.72 MHz Oscillator Circuit:

Hardware modifications to the IC-725's 30.72 MHz Oscillator circuit are:

1.  Remove the stock 30.72 MHz crystal (CR-21 on the "PLL Unit" PCB) and replace it with an Icom CR-64.

2.  Change the DC power to the CR-64's heater pin:

• Lift the leg of L33 closest to C132 (47uF electrolytic).
• Solder LM2904CT-12 voltage regulator to the side of the oscillator shield wall.
• Add input and output capacitors to the LM2940.
• Connect the LM2940's input pin to the lifted-pin of L33.
• Connect the LM2940's output pin to the L33 PCB pad that had held the now-lifted pin.

 3.  Connect a 2-pin female header cable to the LM2940 +12V output and ground (this will supply +12 VDC to the Arduino board).

Notes on the IC-725 30.72 MHz modifications:

The LM2940, being a Low-Dropout (LDO) regulator, has specific requirements regarding the ESR and capacitance of its output capacitor.  For regulator stability this cap must be at least 22 uF and its ESR should be within the ranges shown in the graph, below:

(Refer to the LM2940 datasheet for more details).

The oven of my CR-64 draws about 80 mA when powered with 12V (after it has warmed up).  This current, in addition to the current for the Arduino and Si5351A should be less than 200 mA, so the ESR for the output capacitor should be in the range of 0.1 to 1 ohm to ensure regulator stability.  I used a 22 uF tantalum capacitor from my junkbox whose ESR measured about 200 milliohms (measured on my HP 4274A).

(Note that this tantalum cap is in parallel with the 47 uF aluminum-electrolytic cap (C132) on the radio's PCB.  So the equivalent-ESR will be lower, but not by too much -- Aluminum Electrolytic capacitors typically have significantly higher ESR values.

Never the less, I used my oscilloscope to verify that the output of the LM2940 had no oscillations.

Here's the implementation of the LM2940 circuit.  Note that my IC-725 does not have a shield "cap" for the 30.72 MHz oscillator section.

With the CR-64 powered by the radio's 13.8VDC supply (i.e. the "stock" connection for the CR-64), I discovered that there was some frequency drift (in one direction) during WSPR TX that would then drift back during RX, as shown by the screen captures, below.  (The signal I'm monitoring is generated via my HP 3335A generator (with a short wire as an antenna at its output), set to 14.0972 Mhz (= 14.0956 MHz + 1600Hz) :

(Note that my 14.0972 MHz signal isn't exactly on WSPR's 1600 Hz frequency.  This offset is due to errors in the 30.72 MHz Oscillator frequency and in the 9.013 MHz BFO USB frequency.)

With the added LM2940, the CR-64's heater is powered with a DC voltage (+12V) that is constant irrespective of mode. Drift is now much better:

 

Fan Control Modifications (and Frequency Drift due to Rapid Temperature Change):

(Note:  I changed the Fan to be "always ON" (rather than only being ON during TX) in an attempt to improve frequency stability when I was using a VCTCXO to generate 30.72 MHz.  I kept this modification with the CR-64, reasoning that minimizing temperature deltas between TX and RX modes would be a good thing even with the CR-64.)

Below is a discussion of  this modification...

I had noticed that the metal heat-sink to which the 8V regulator and the speaker amplifier are attached can get very warm.  For example, after allowing some time for its temperature to stabilize, it measured at 72 degrees C (ambient was around 30 degrees C).  

That's hotter than I'd like to see it, given that the electrolytic capacitors next to it are only rated to 85 degrees C.  I'd personally prefer that there be more of a margin, for reliability purposes.

With this amount of heat being generated, the newly-added temperature-compensated crystal oscillators also heat up.

Normally, the heating of the TCXOs would not be an issue because they are, after all, temperature compensated.  But if I transmitted a CW signal, I noticed that over the course of the first minute there could be a drift in frequency of, for example, 26 Hz at 28.5 MHz.

This drift-during-TX was lower at lower frequencies (e.g. 9 Hz at 3.5 MHz), implying that the problem was with the 30.72 MHz VCTCXO rather than the BFO oscillator.

What could be causing it?

Everything I measured seemed to be fine, and then I realized that one difference between TX and RX was that, during TX, the PA fan would turn on, and then it would turn off  when the transceiver went back to receive.

Could the fan be cooling the VCTCXO too rapidly for its temperature-compensating circuitry to compensate for the abrupt change in temperature?  

Unfortunately, the Connor-Winfield VCTCXO has no specification for "Maximum Temperature Rate of Change," but, searching Digikey, I did find a VCTCXO (IQD LFTVXO076345REEL) that included this specification.  The IQD VCTCXO's  "temperature rate-of-change" is spec'd at 1 degree C per minute, max!

I am sure that the temperature of the VCTCXO changes more than one degree in one minute when the IC-725's fan kicks on (or off).  If the Connor-Winfield VCTCXO has a similar limitation, then this would be the most likely cause of the significant frequency-change-during-Transmit that I was seeing.

Of course, the change in frequency could also be due to the VCTCXO's Frequency Control voltage (10-turn pot plus voltage reference) changing with temperature, but when I checked this with my voltmeter (HP 3478A), I did not see any appreciable voltage change.

As an experiment, I modified the IC-725 fan-controller circuitry by shorting Q7's emitter to its collector.  This mod keeps the fan ON whenever the radio is ON :

With this modification added, the frequency shifted by 0.9 Hz during 1 minute of key-down CW transmission on 28.5 MHz.  Much better that the 26 Hz of drift I'd originally measured.

And as an added benefit, the "always-on" fan should help lower the temperature of the hot 7808 8-volt regulator.

Discarded 30.72 MHz Oscillator Options:

Let's look at the two 30.72 MHz oscillator options that I discared:

1.  Using the Si5351A to generate 30.72 MHz.

Generating 30.72 MHz using the Si5351A and its 25 MHz reference is a straight-forward task, given  that the Si5351A has two internal oscillators and three outputs.

But fractional-division is going to occur somehow, and thus some amount of spurs will be generated.  (I recommend that it be within the PLL feedback loop and not at the Si5351A's output -- set that divisor to be an even integer.  I used the Si5351A's CLK2 output and I set its output divisor to 34).

Here's a look at the output spurs...

Si5351 30.72 MHz Oscillator Spurs:

Spurs on the 30.72 MHz oscillator could be a problem because, given the wide bandwidth of the input signal into the mixer assigned to it, spurs could cause undesirable mixing products to appear in the RX passband.

Close in spurs within +/- 10 KHz look great (20 KHz span):

If the span is increased to 1 MHz, spurs become more evident:

And even more so if the span is set from 1 to 100 MHz:

The spurs surrounding the 30.72 MHz fundamental could be a problem.  When I connected an HP 8640B signal generator to the input of the transceiver, set the transceiver's receive frequency to 10.15 MHz and then started tuning the 8640B away from this frequency in either direction, I could hear spur products.

2.  Using a separate VCTCXO for 30.72 MHz:

I had initially installed a TCXO to generate the 30.72 MHz reference-oscillator signal, but after I installed it I discovered it was too far off frequency, and so I changed the design to use a voltage-controlled TCXO (VCTCXO) with a 10-turn pot to control the oscillator's frequency.

Here are the spectrum plots of the Conner-Winfield T604-30.72M VCTCXO that I used.

First, 5 KHz span:

Next, 1 MHz span (I skipped 20 KHz span):

And finally, the span from 1 to 100 MHz:

You can see that, although the amplitude of the Conner-Winfield VCTCXO is about 1 dB lower than the level of the 30.72 MHz signal when generated with the Si5351, its spectral content is much improved.

To check stability over time, I used WSJT-X to monitor a signal at 14097.2 MHz (created with my HP 3335A generator and with a short wire as an antenna at its output).

Over time (e.g. 24 hours) I would see a shift of about +/- 10 Hz in this frequency.  Not what I was expecting to see!

There were a number of potential causes -- VCTCXO short-term "aging", or fan speed variation between TX and RX, due to a slight lowering of the 13.8 VDC to the radio during TX (the fan runs a bit slower during TX).  Or the radio's interior warming up slightly over time due to the hot 7808 8-volt regulator.

Whatever the reason, trying to find the cause and fix it looked like it was going to be a headache, so I opted for my third choice -- use an Icom CR-64 High-Stability Crystal Unit.

By the way, for completeness, here's the schematic of the VCTCXO version:

And here are the PCB mods:

NOTE THAT THESE MODIFICATIONS ARE NOT BEING USED!

Daughter-board Build and Installation:

Below is my build of the circuit shown in the schematic.  Note that the photos below are of the VCTCXO version of the 30.72 MHz oscillator.  The current design looks the same, except (1) the 30.72 MHz SMA cable has been removed, and (2) the wire from the 3.3V regulator's output to the VCTCXO's power pin has been removed.

The board mounts behind the front panel of the IC-725 using the three screw locations shown below:

And here is the board, mounted on the front-panel assembly:

(Yellow arrows point to the three mounting screws).

And here's how it looks with the front-panel assembly mounted to the radio:

Again, I'll point out, because the VCTCXO (also-known-as TCVCXO) is no longer used, the potentiometer is no longer used, too.

Arduino Code:

The Arduino code can be found on my GitHub site, here:  

https://github.com/k6jca/IC-725_Si5351/tree/Rev_200929

A few words regarding frequency generation with the Si5351:

My code is based, in part, on the uBitx Si5351 routines written by Jerry Gaffke, KE7ER (the uBitx is a popular QRP transceiver).  

However, the uBitx code changes the uBitx radio's frequency by changing the Output Divider of the Si5351.  As I explain, below, to minimize spurs for my application, I found it better to keep the Output Divider constant (and set to an even integer for, in theory, a symmetric waveform).  And for this reason my code diverges from the original uBitx frequency-control philosophy.

But more on this in a bit.  First, let's look at the Si5351 architecture...

The Si5351 has two sets of registers that control two fractional dividers.

One fractional divider controls the internal PLL's output frequency (i.e. the high-frequency intermediate clock, which I set to around 703 MHz for this application).  The other fractional divider divides this internal high-frequency intermediate clock down to the final clock frequency (which is around 9 MHz, for the IC-725 BFO).

To limit output spurs, I recommend setting the Output Fractional Divider to an even integer (e.g. 28 in this application).  An even-integer does not prevent spurs from occurring, but it should reduce their number, as the following plots show.  And, an even integer divider would equate to a symmetric output waveform, assuming the input to the divider was unchanging.

The image below shows the output of the Si5351 when it has been programmed to generate the CW (RX) BFO frequency, but with the High-Frequency Intermediate Clock set to 875 and the Output Fractional Divider set to a non-integer value.  Note the spur that is only 200 Hz from the main BFO frequency.

On the other hand, with the High-Frequency Intermediate Clock set to 702.7644 MHz and the Output Fractional Divider set to 28, the spurs for the same 5 KHz span are:

Certainly an improvement!

The following two plots show the spurs over a span of 20 KHz.  The first plot is the spectrum of the non-integer Output Fractional Divider, while the second plot is of the even-integer Output Fractional Divider.

The second one (even-integer Output Fractional Divider) still looks better.

And finally, the following two plots are the spectrums from 2 MHz to 30 Mhz.  Again, note that the even-integer Output Fractional Divider setting (second plot) looks better.

A final comment...

Although there are still quite a few spurs (apart from  the expected harmonics of the BFO frequency) over the 2-30 MHz range, in my opinion these are not too important given that the BFO mixing is done after the IF filter (which has, for SSB, a 2.1 KHz bandwidth). 

Therefore, if the spurs lie outside this filter bandwidth, then they are mixing with greatly reduced signal levels (or the noise floor) in the filter's stop-band and probably don't add much to the overall system noise.

On the other hand, spurs that lie within the filter's passband (or on its slopes) could add to interference, and thus should be avoided.  For this reason, I recommend that the Output Fractional Divider be set to an even-integer divider.

A Note Regarding the Use of an External Oscillator with the Silicon Labs Si5351A-B-GT: 

Silicon Labs only specifies the Si5351A-B-GT part to be used with a crystal as its frequency reference.  It does not specify the part for use with an external oscillator such as the TCXO I used -- it is only spec'd for use with a 25 or 27 MHz crystal as its Frequency Reference.

On the other hand, the Si5351C part is spec'd for use with an external oscillator as its frequency reference.  However, the Si5351C is only available in an QFN package, and  I have not yet found anyone who supplies a pre-fabricated development board for this part.

Therefore, for the design in this blog post, I am using a pre-fabricated development board (from AdaFruit) with an Si5351A and associated circuitry already installed upon it, and I have modified it to drive the Si5351A's crystal input with an external oscillator (TCXO) whose output has been modified to mimic the Si5351A's crystal-input signal.

This thread on the Silicon Labs forum highlights the issue of driving the crystal-only chip with an external oscillator:   https://www.silabs.com/community/timing/forum.topic.html/si5351_msop10_packag-wocK

Note the statement that the part is only spec'd for use with a crystal.

But does this mean that an external oscillator can't be used?  Perhaps not, as pointed out here. 

But speaking as an engineer with 40 years of design experience, I would never intentionally design a product that did not follow a manufacturer's specifications -- doing so can lead to a product-support nightmare (for example, if the part's manufacturing processes changes and the part begins failing in my product) which would lead back to only one and only one root cause: me, and not the part manufacturer.

For this "one-off" design, though, I decided to take the risk.  Why not make the TCXO's signal look similar to a crystal's signal at the XA pin of the Si5351A?  Maybe this would minimize my risk of using the part outside of Silicon Lab's specs.

First thing to do -- measure the signals at the Si5351's XA and XB pins:

XA pin with xtal:  Amplitude = 570 mVPP, DC Offset = 420 mV

XB pin with xtal:  Amplitude = 1.04 VPP.

Next, drive the XA pin with the TCXO (via a voltage divider, one-pole RC filter, and capacitively coupled):

XA pin with TCXO:  Amplitude = 530 mVPP, DC Offset = 450 mV.

XB pin with xtal:  Amplitude = 1.0 VPP.

Summary:  the Si5351A XA and XB pin waveforms, with the TCXO, are similar to (but not exactly matching) those waveforms created with the original 25 MHz crystal.

One difference: note that the XA pin's waveform with the TCXO is similar to a square wave (but filtered by both the scope's response as well as the one-pole RC filter at the TCXO's output. Therefore the 25 MHz fundamental of the square wave will have a larger amplitude than what is shown, above,  (probably on the order of 530mVPP*4/pi (the latter the ratio of fundamental amplitude to square-wave amplitude), or 670 mVpp).

So...the TCXO signal is not exactly the same at pin XA, but it is closer than if I were driving pin XA (capacitively coupled) with the 3.3Vpp square-wave from the TCXO's output.  

For my own use, this seems to be good enough.  But if I wanted to improve upon XA's waveform, additional filtering (and a change to the voltage-division ratio) could be applied to the TCXO waveform to bring it closer to that of the crystal-generated waveform.

Other Bits of Flooby-dust:

8-volt Regulator:

As I mentioned above (in the fan section) the 7808 8-volt linear regulator and its heatsink get hot.  It sources about 0.25 amps (measured in CW mode), which means, given an input voltage of 13.8 volts, it's dissipating about 1.5 Watts of power.

I thought I would try replacing it with a 5V switching regulator (buck converter, found on Amazon) that I would convert to 8 volts:

The code on the IC package is AGCE, which corresponds to an MP2315 500-KHz switching regulator, and the board's circuit is very similar to the schematic (shown below) found on the Monolithic Power Systems website.  (There are a few extra parts on the actual board, such as LED, TVS on output, etc., but, per the 3-letter codes on the resistor packages, the resistor values are almost all the same (note that R2 is 7.32K on the board rather than the schematic's 7.5K).

I removed the unneeded USB connector and cut off the unused PCB area to reduce its size.  I then removed the TVS 5V over-voltage protection (connected to the board's output voltage node) and paralleled the on-board 7.32K resistor with 10.7K ohms to get about 8 VDC out. 

But when I installed this board in the radio in lieu of the original 7808 linear regulator, I discovered that it generated noticeable EMI at 7.135 MHz (I only checked the 40-meter band), and so I removed it and reinstalled the original 8V linear regulator.

30.72 MHz Signal Level:

(Note:  this discussion is for the now-discarded design  using the VCTCXO)

When I started making my modifications I neglected to measure the original signal level of the 30.72 MHz oscillator (and I was too lazy to go through the effort of unhooking the cables from the board, removing it, resoldering the crystal onto it, then reinstalling the board and reconnecting all of the cables to it).

Instead, to determine what would be an adequate oscillator signal level, I used an HP 3335A signal generator set to 30.72 MHz and connected it to the SMA I had installed at the IC-725's for injecting a 30.72 MHz clock.

While listening to the audio of a received CW signal, I adjusted the signal generator's output from a low level until, to my ears, it seemed that the audio output level was no longer increasing.

This plateauing of audio output occurred when the 3335A's output was around +9 to +10 dBm.  

Measuring (using my TDS 350 oscilloscope) the signal at  J4 of the IC-725 (with P4 inserted) and +9 dBm of drive from the HP 3335A, the signal level was 740 mVpp (248 mVRMS using the TDS 350's RMS calculator).

When driven with the VCTCXO, the voltage at J4 measured to be 780 mVpp (267 mVRMS), putting it within my target goal of being equivalent to the 3335A output of +9 to +10 dBm.

30.72 MHz secondary oscillations:

(Note:  this discussion is for the now-discarded design  using the VCTCXO)

The VCTCXO drives the base of the oscillator's transistor in lieu of the 30.72 MHz that was originally in the circuit (I did this so that I would not need to remove many parts).  I also removed the base-emitter capacitor, turning the circuit into an emitter-follower for the Si5351 signal.

While testing with the Si5351 generating the 30.72 MHz, I tried putting different filters (from Mini-Circuits) in series with the Si5351's output in an attempt to lower the spur content of the 30.72 MHz signal.

But, surprising to me, these filters had an undesired effect -- they caused the IC-725's oscillator to start oscillating again, but at a different frequency.  And so there were two frequencies at the output of this oscillator -- the Si5351 frequency and a different (but fairly close) frequency.

What was going on?

Although I had removed the original crystal that was the frequency-determining component of the IC-725's oscillator, by connecting a filter in series with the Si5351's output to the base of the IC-725's oscillator transistor I was connecting a new resonant circuit (composed of the filter's inductors and capacitors) to the transistor's base, with the other side of the resonant circuit effectively connected to a low-impedance point (i.e. the output of the Si5351).

And even though I had removed the original base-emitter capacitor, the original oscillator circuit was now seeing the filter as a new resonant circuit in lieu of the original crystal and oscillating at the frequency determined by the filter's components.

In addition, the circuit's transistor was still acting as an emitter follower for the Si5351 signal.  So it was self-oscillating as well as acting as a buffer for the Si5351-generated signal.

The fix -- remove the filter (and stop the unwanted oscillations) and replace the Si5351 with the VCTCXO (eliminate spurs around the 30.72 MHz signal).

Radio Stability with these Modifications:

At the moment the IC-725 has been running WSPR for a day and a half and I am satisfied with the improved stability of the IC-725.  However, while monitoring my HP 3335A-generated 14.0972 MHz "marker signal" on WSPR, I have noticed a long-term drift of roughly 10 Hz over 24 hours.

And during this period, when the radio does drift, it's usually fairly quickly -- i.e. several Hz within, say, 2 minutes, and then it seems to stabilize and not change (or change imperceptibly) for a long time.

I haven't investigated if this drift is due to the CR-64 or due to the TCXO driving the Si5351.  This would be done by also checking the frequency offset of a marker on, say, on the 10-meter WSPR band -- if the drift on 10 meters and 20 meters is the same amount (in terms of Hz), then the BFO is dominating drift.  But if they are different, then the 30.72 MHz oscillator is dominating drift.

For the moment I'm assuming that it's the CR-64's crystal stabilizing over time, in the same way that an OCXO might take a few days to stabilize (as I've seen with my GPSDO's HP OCXO, when powered-up from a cold-start).

Future Modifications?

Because my 30.72 MHz VCTCXO modification is still in place within the radio (but with no power applied to it because I am now using the CR-64), one future modification might be to added a PLL circuit (such as the Analog Devices ADF4001 "Integer-N PLL" IC, plus filter) and lock the 30.72 MHz oscillator to my lab GPSDO frequency reference (for my Arduino-based GPSDO, see here).

Other Links of Interest:

Modifying an IC-751A for Digital Modes:  https://squashpractice.com/2017/03/06/modifications-to-the-ic-751a-for-the-digital-modes/ (Note that the mods include replacing the stock 30.72 MHz oscillator with an OCXO and small switching power-supply module -- this is a mod I would have tried if I hadn't found a CR-64 (on eBay), but I would be concerned that EMI from the switcher might interfere with the radio).

Insulating the IC-746A's oscillator with cotton-wool balls:  https://www.chris.org/Modifications/IC-746-almost-free-tcxo.html

Standard Caveat:

I might have made a mistake in my designs, equations, schematics, models, etc. If anything looks confusing or wrong to you, please feel free to comment below or send me an email.

Also, I will note:

This design and any associated information is distributed in the hope that it will be useful, but WITHOUT ANY WARRANTY; without even the implied warranty of MERCHANTABILITY or FITNESS FOR A PARTICULAR PURPOSE.

Understanding the Basic RF Field Strength Meter

 

(Radio Shack SWR/Field-Strength Meter, purchased circa 1970 when I was a young Novice (original 100uA meter replaced with generic meter during the early 1970s))

A friend of mine who was building a field-strength meter as a tuning-aid for his magnetic-loop antenna asked my opinion of the following schematic:

I took a quick look at it and told him that I thought that diode D2 was not needed (thinking that if D1 charged up C1, then D2 would have no role in the determining C1's voltage -- after all, if D2 were conducting, then D1 would be back-biased and C1's voltage would remain unchanged).

Well, I was wrong.

My friend built the field-strength meter and quickly determined that the second diode was needed.  Without it, the meter would always read zero.

Why was the second diode required?  Clearly, I needed to take a closer look...

Creating a Model for Analysis:

Field-strength meters, given that they use a whip antenna, can be thought of as an electric-field (E-field) detector, rather than a magnetic field detector.

When used, the typical field-strength meter might be sitting on a table-top or mounted on a wall or even hand-held.  That is, they are often not directly connected to ground:

What is the signal path between the transmitter and the field-strength meter when the field-strength meter is not directly connected to ground?

The image below shows a path that is not the signal path, given a meter being used as an E-field detector.  (It would be a valid path if the circuit were used as a magnetic field probe, but in such an application the whip antenna would not be needed and you could replace one diode with a short).

Continuing with the field-strength meter as an E-field detector analogy, for purposes of analysis we can picture the field-strength meter as being capacitively coupled (E-field coupled) to the transmitter and to ground via the three paths shown in the image, below:

And again, for purposes of analysis, let's assume that one of the two legs capacitively coupling the meter's terminals to ground has more capacitance than the other leg, so that it dominates. 

Note:  It doesn't matter which leg dominates, the meter works in either case (as can be shown with LTSpice simulations).  In fact, it also works if the capacitances of the two legs are equal.

But for simplicity, let's treat the meter as having the following two capacitively coupled paths:

Because the two capacitors in the signal path (antenna to antenna, meter to ground) are in series with the field-strength meter, for analysis purposes I can combine these two series-capacitors into an equivalent single capacitor.  Also, for simulation I'll remove the capacitor across the meter and replace the meter with a resistor.  Thus, any current flowing through diode D2 also flows through the resistor. 

Here's the LTSpice circuit:

The driving signal is a sinusoidal voltage at 10 MHz.  The image below shows the simulation results. 

You can see that the voltage across the resistor R1 (i.e. through the meter) is a pulsating positive DC voltage.  Therefore, the meter would deflect from 0 by an amount related to the RMS level of this pulsating waveform.

Note, too, that current flows through C3 during the entire cycle of the sinusoidal source. 

The simulation below is of the same circuit, but it shows the voltage across the capacitor C3 as well as the diode-clamping effect at node B:

And for completeness, below are two simulations.  The first has the capacitance in the other leg dominate and the second has both "leg" capacitances equal:

Note that for the "equal leg" simulation the voltage across R1 looks more like a full-wave rectifier's output.

Circuit with One Diode:

But what happens if I remove a diode?

Below is the LTSPICE circuit (with the two series-capacitors in the loop combined into one):

And here is its simulation:

Note that with the first positive transition of the driving source's sine-wave, capacitor C3 charges up to a value essentially equal to V1(pk) - Vf(diode) (assuming that the voltage drop across R1 is much smaller than Vf(diode).

During the positive portion of the source's cycle, the circuit's loop equation is:

V1 -VC3 - Vdiode - VR1 = 0

Or,

VR1 = V1 - VC3 - Vdiode

Given that VC3 = V1(pk) - Vf(diode), then the diode is never be forward biased into conduction during the sine wave's positive cycle and thus no current would flow through the meter.

During the negative portion of the source's cycle, the diode is always back-biased.  Thus no current flows and the meter never deflects.

And so, if the circuit only has one diode, no current flows through the meter (apart from the initial transient) and its reading is always zero.

(Note:  as a general design rule this means that, when capacitively coupling a diode-detector to a voltage source, you should use two diodes and not one).

Other comments:

If the capacitors are large enough that their impedances are low at the frequency of operation, then the circuit is similar to that of a classic voltage doubler.

Here's a classic voltage doubler:

And here's the FSM with its "implied" capacitive couplings set to large values of capacitance:

Standard Caveat:

I might have made a mistake in my designs, equations, schematics, models, etc. If anything looks confusing or wrong to you, please feel free to comment below or send me an email.

Also, I will note:

This design and any associated information is distributed in the hope that it will be useful, but WITHOUT ANY WARRANTY; without even the implied warranty of MERCHANTABILITY or FITNESS FOR A PARTICULAR PURPOSE.